Load control device for high-efficiency loads

ABSTRACT

A two-wire load control device (such as, a dimmer switch) for controlling the amount of power delivered from an AC power source to an electrical load (such as, a high-efficiency lighting load) includes a thyristor coupled between the source and the load, a gate coupling circuit coupled between a first main load terminal and the gate of the thyristor, and a control circuit coupled to a control input of the gate coupling circuit. The control circuit generates a drive voltage for causing the gate coupling circuit to conduct a gate current to thus render the thyristor conductive at a firing time during a half cycle of the AC power source, and to allow the gate coupling circuit to conduct the gate current at any time from the firing time through approximately the remainder of the half cycle, where the gate coupling circuit conducts approximately no net average current to render and maintain the thyristor conductive.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.15/852,328, filed on Dec. 22, 2017, which is a continuation of U.S.patent application Ser. No. 15/130,288, filed Apr. 15, 2016, now U.S.Pat. No. 9,853,561, issued Dec. 26, 2017, entitled LOAD CONTROL DEVICEFOR HIGH-EFFICIENCY LOADS, which is a continuation of U.S. patentapplication Ser. No. 14/581,115, filed Dec. 23, 2014, now U.S. Pat. No.9,343,997, issued May 17, 2016, entitled LOAD CONTROL DEVICE FORHIGH-EFFICIENCY LOADS, which is a continuation of U.S. patentapplication Ser. No. 13/458,324, filed Apr. 27, 2012, now U.S. Pat. No.8,957,662, issued Feb. 17, 2015, entitled LOAD CONTROL DEVICE FORHIGH-EFFICIENCY LOADS which is a continuation-in-part ofcommonly-assigned U.S. patent application Ser. No. 13/232,344, filedSep. 14, 2011, now U.S. Pat. No. 8,698,408, issued Apr. 15, 2014,entitled TWO-WIRE DIMMER SWITCH FOR LOW-POWER LOADS, which is acontinuation-in-part of commonly-assigned U.S. patent application Ser.No. 12/952,920, filed Nov. 23, 2010, now U.S. Pat. No. 8,664,881, issuedMar. 4, 2014, entitled TWO-WIRE DIMMER SWITCH FOR LOW-POWER LOADS, whichclaims priority from U.S. Provisional Patent Application Ser. No.61/264,528, filed Nov. 25, 2009, and U.S. Provisional Patent ApplicationSer. No. 61/333,050, filed May 10, 2010, both entitled TWO-WIRE ANALOGDIMMER SWITCH FOR LOW-POWER LOADS, the entire disclosures of which arehereby incorporated by reference.

BACKGROUND OF THE INVENTION Field of the Invention

The present invention relates to load control devices for controllingthe amount of power delivered to an electrical load, and moreparticularly, to a two-wire analog dimmer switch for controlling theintensity of a low-power lighting load, such as a light-emitting diode(LED) light source having an LED driver circuit or a fluorescent lamphaving an electronic dimming ballast.

Description of the Related Art

Prior art two-wire dimmer switches are coupled in series electricalconnection between an alternating-current (AC) power source and alighting load for controlling the amount of power delivered from the ACpower source to the lighting load. A two-wire wall-mounted dimmer switchis adapted to be mounted to a standard electrical wallbox and comprisestwo load terminals: a hot terminal adapted to be coupled to the hot sideof the AC power source and a dimmed hot terminal adapted to be coupledto the lighting load. In other words, the two-wire dimmer switch doesnot require a connection to the neutral side of the AC power source(i.e., the load control device is a “two-wire” device). Prior art“three-way” dimmer switches may be used in three-way lighting systemsand comprise at least three load terminals, but do not require aconnection to the neutral side of the AC power source.

The dimmer switch typically comprises a bidirectional semiconductorswitch, e.g., a thryristor (such as a triac) or two field-effecttransistors (FETs) in anti-series connection. The bidirectionalsemiconductor switch is coupled in series between the AC power sourceand the load and is controlled to be conductive and non-conductive forportions of a half cycle of the AC power source to thus control theamount of power delivered to the electrical load. Generally, dimmerswitches use either a forward phase-control dimming technique or areverse phase-control dimming technique in order to control when thebidirectional semiconductor switch is rendered conductive andnon-conductive to thus control the power delivered to the load. Thedimmer switch may comprise a toggle actuator for turning the lightingload on and off and an intensity adjustment actuator for adjusting theintensity of the lighting load. Examples of prior art dimmer switchesare described in greater detail is commonly-assigned U.S. Pat. No.5,248,919, issued Sep. 29, 1993, entitled LIGHTING CONTROL DEVICE; U.S.Pat. No. 6,969,959, issued Nov. 29, 2005, entitled ELECTRONIC CONTROLSYSTEMS AND METHODS; and U.S. Pat. No. 7,687,940, issued Mar. 30, 2010,entitled DIMMER SWITCH FOR USE WITH LIGHTING CIRCUITS HAVING THREE-WAYSWITCHES, the entire disclosures of which are hereby incorporated byreference.

With forward phase-control dimming, the bidirectional semiconductorswitch is rendered conductive at some point within each AC line voltagehalf cycle and remains conductive until approximately the next voltagezero-crossing, such that the bidirectional semiconductor switch isconductive for a conduction time each half cycle. A zero-crossing isdefined as the time at which the AC line voltage transitions frompositive to negative polarity, or from negative to positive polarity, atthe beginning of each half cycle. Forward phase-control dimming is oftenused to control energy delivered to a resistive or inductive load, whichmay include, for example, an incandescent lamp or a magnetic low-voltagetransformer. The bidirectional semiconductor switch of a forwardphase-control dimmer switch is typically implemented as a thyristor,such as a triac or two silicon-controlled rectifiers (SCRs) coupled inanti-parallel connection, since a thyristor becomes non-conductive whenthe magnitude of the current conducted through the thyristor decreasesto approximately zero amps.

Many forward phase-control dimmers include analog control circuits (suchas timing circuits) for controlling when the thyristor is renderedconductive each half cycle of the AC power source. The analog controlcircuit typically comprises a potentiometer, which may be adjusted inresponse to a user input provided from, for example, a linear slidercontrol or a rotary knob in order to control the amount of powerdelivered to the lighting load. The analog control circuit is typicallycoupled in parallel with the thyristor and conducts a small timingcurrent through the lighting load when the thyristor is non-conductive.The magnitude of the timing current is small enough such that thecontrolled lighting load is not illuminated to a level that isperceptible to the human eye when the lighting load is off.

Thyristors are typically characterized by a rated latching current and arated holding current, and comprise two main load terminals and acontrol terminal (i.e., a gate). The current conducted through the mainterminals of the thyristor must exceed the latching current for thethyristor to become fully conductive. In addition, the current conductedthrough the main terminals of the thyristor must remain above theholding current for the thyristor to remain in full conduction. Since anincandescent lamp is a resistive lighting load, a typical forwardphase-control dimmer switch is operable to conduct enough currentthrough the incandescent lamp to exceed the rated latching and holdingcurrents of the thyristor if the impedance of the incandescent lamp islow enough. Therefore, prior art forward phase-control dimmer switchesare typically rated to operate appropriately with lighting loads havinga power rating above a minimum power rating (e.g., approximately 40 W)to guarantee that the thyristor will be able to latch and remainedlatched when dimming the lighting load.

Some prior art dimmer switches have included two triacs coupled togetherto overcome some of the problems related to the rated latching andholding currents of triacs as described in greater detail incommonly-assigned U.S. Pat. No. 4,954,768, issued Sep. 4, 1990, entitledTWO WIRE LOW VOLTAGE DIMMER. Such a prior art dimmer switch may comprisea first triac characterized by a low power rating and low latching andholding currents, and a second triac characterized by a high powerrating and high latching and holding currents. The main load terminalsof the first triac are coupled between one of the main load terminalsand the gate of the second triac. In addition, a resistor is coupledbetween the other main load terminal and the gate of the second triac.If the magnitude of the load current is small, the first triac isrendered conductive when a pulse of current is conducted through thegate and remains latched until the magnitude of the load current dropsbelow the holding current of the first triac (e.g., at the end of a halfcycle). If the magnitude of the load current is large, the first triacconducts a pulse of the gate current through the gate of the secondtriac to render the second triac conductive and the second triacconducts the load current. Since the voltage across the first triacdrops to approximately zero volts when the second triac is conductive,the first triac becomes non-conductive after the second triac isrendered conductive. The second triac remains conductive until themagnitude of the load current drops below the holding current of thesecond triac (e.g., at the end of a half cycle).

When using reverse phase-control dimming, the bidirectionalsemiconductor switch is rendered conductive at the zero-crossing of theAC line voltage and rendered non-conductive at some point within eachhalf cycle of the AC line voltage, such that the bidirectionalsemiconductor switch is conductive for a conduction time each halfcycle. Reverse phase-control dimming is often used to control energy toa capacitive load, which may include, for example, an electroniclow-voltage transformer. Since the bidirectional semiconductor switchmust be rendered conductive at the beginning of the half cycle, and mustbe able to be rendered non-conductive within the half cycle, reversephase-control dimming requires that the dimmer switch have two FETs inanti-serial connection, or the like. A FET is operable to be renderedconductive and to remain conductive independent of the magnitude of thecurrent conducted through the FET. In other words, a FET is not limitedby a rated latching or holding current as is a thyristor. However, priorart reverse phase-control dimmer switches have either required neutralconnections and/or advanced control circuits (such as microprocessors)for controlling the operation of the FETs. In order to power amicroprocessor, the dimmer switch must also comprise a power supply,which is typically coupled in parallel with the FETs. These advancedcontrol circuits and power supplies add to the cost of prior artFET-based reverse phase-control dimmer switches (as compared to analogforward phase-control dimmer switches).

Further, in order to properly charge, the power supply of such atwo-wire dimmer switch must develop an amount of voltage across thepower supply and must conduct a charging current from the AC powersource through the electrical load, in many instances even when thelighting load is off. If the power rating of the lighting load is toolow, the charging current conducted by the power supply through thelighting load may be great enough to cause the lighting load toilluminate to a level that is perceptible to the human eye when thelighting load is off. Therefore, prior art FET-based reversephase-control dimmer switches are typically rated to operateappropriately with lighting loads having a power rating above a minimumpower rating to guarantee that the lighting load does not illuminate toa level that is perceptible to the human eye due to the power supplycurrent when the lighting load is off. Some prior art load controldevices, have included power supplies that only develop small voltagesand draw small currents when charging, such that the minimum powerrating of a controlling lighting load may be as low as 10 W. An exampleof such a power supply is described in greater detail incommonly-assigned U.S. patent application Ser. No. 12/751,324, filedMar. 31, 2010, entitled SMART ELECTRONIC SWITCH FOR LOW-POWER LOADS, theentire disclosure of which is hereby incorporated by reference.

Nevertheless, it is desirable to be able to control the amount of powerto electrical loads having power rating lower than those able to becontrolled by the prior art forward and reverse phase-control dimmerswitches. In order to save energy, high-efficiency lighting loads, suchas, for example, compact fluorescent lamps (CFLs) and light-emittingdiode (LED) light sources, are being used in place of or as replacementsfor conventional incandescent or halogen lamps. High-efficiency lightsources typically consume less power and provide longer operationallives as compared to incandescent and halogen lamps. In order toilluminate properly, a load regulation device (e.g., such as anelectronic dimming ballast or an LED driver) must be coupled between theAC power source and the respective high-efficiency light source (i.e.,the compact fluorescent lamp or the LED light source) for regulating thepower supplied to the high-efficiency light source.

A dimmer switch controlling a high-efficiency light source may becoupled in series between the AC power source and the load controldevice for the high-efficiency light source. Some high-efficiencylighting loads are integrally housed with the load regulation devices ina single enclosure. Such an enclosure may have a screw-in base thatallows for mechanical attachment to standard Edison sockets and provideelectrical connections to the neutral side of the AC power source andeither the hot side of the AC power source or the dimmed-hot terminal ofthe dimmer switch (e.g., for receipt of the phase-control voltage). Theload regulation circuit is operable to control the intensity of thehigh-efficiency light source to the desired intensity in response to theconduction time of the bidirectional semiconductor switch of the dimmerswitch.

However, the load regulation devices for the high-efficiency lightsources may have high input impedances or input impedances that vary inmagnitude throughout a half cycle. Therefore, when a prior-art forwardphase-control dimmer switch is coupled between the AC power source andthe load regulation device for the high-efficiency light source, theload control device may not be able to conduct enough current to exceedthe rated latching and/or holding currents of the thyristor. Inaddition, when a prior-art reverse phase-control dimmer switch iscoupled between the AC power source and the load regulation device, themagnitude of the charging current of the power supply may be greatenough to cause the load regulation device to illuminate the controlledhigh-efficiency light source to a level that is perceptible by the humaneye when the light source should be off.

The impedance characteristics of the load regulation device maynegatively affect the magnitude of the phase-control voltage received bythe load regulation device, such that the conduction time of thereceived phase-control voltage is different from the actually conductiontime of the bidirectional semiconductor switch of the dimmer switch(e.g., if the load regulation device has a capacitive impedance).Therefore, the load regulation device may control the intensity of thehigh-efficiency light source to an intensity that is different than thedesired intensity as directed by the dimmer switch. In addition, thecharging current of the power supply of the dimmer switch may build upcharge at the input of a load regulation device having a capacitiveinput impedance, thus negatively affecting the low-end intensity thatmay be achieved.

Therefore, there exists a need for a two-wire load control device thatmay be coupled between an AC power source and a load regulation devicefor a high-efficiency light source and is able to properly control theintensity of the high-efficiency light source.

SUMMARY OF THE INVENTION

The present invention provides a “two-wire” load control device, such asa dimmer switch, for low-power lighting loads, such as a light-emittingdiode (LED) lamp having an LED driver circuit or a compact fluorescentlamp having an electronic dimming ballast. The dimmer switch is able toappropriately control a wider variety of types of lamps, as well assimpler and less expensive lamps, than the prior art dimmer switches.The dimmer switch provides a pure phase-cut waveform to the lampindependent of the specific characteristics of the lamp, which leads toan improved operation of the lamp. The dimmer switch has an improvednoise immunity and works well in a larger variety of installations sincethe dimmer switch has a reduced susceptibility to cross talk that may begenerated by other dimmer switches on the electrical circuit. The dimmerswitch may be a “smart” dimmer switch for providing advanced featuresand functionality, such as an advanced programming mode for adjustingthe operating characteristics of the dimmer switch to improve theperformance of the dimmer switch when working with the LED and CFLlighting loads.

According to an embodiment of the present invention, a load controldevice for controlling the amount of power delivered from an AC powersource to an electrical load comprises a thyristor, a gate couplingcircuit, and a control circuit. The thyristor has first and second mainload terminals adapted to be coupled in series electrical connectionbetween the AC power source and the electrical load for conducting aload current from the AC power source to the electrical load, and a gatethat conducts a gate current to render the thyristor conductive and ischaracterized by a rated holding current. The gate coupling circuit iscoupled to conduct the gate current through the gate of the thyristor.The control circuit is operable to render the controllable switchingcircuit conductive and to control the gate coupling circuit to conductthe gate current to thus render the thyristor conductive at a firingtime during a half cycle of the AC power source. The control circuitcontinues to control the gate coupling circuit, such that the gatecoupling circuit is able to conduct the gate current again after thefiring time. The gate coupling circuit is prevented from conducting thegate current too close to the end of the half-cycle to prevent the triacfrom being rendered conductive at the beginning of the next half-cycle.The gate coupling circuit is also operable to conduct the load current,such that the combination of the thyristor and the gate coupling circuitare operable to conduct the load current through the load independent ofthe rated holding current of the thyristor.

The load control device may also comprise a controllable switchingcircuit coupled between the gate coupling circuit and the gate of thethyristor for conducting the gate current when the controllableswitching circuit is conductive. The control circuit is operable torender the controllable switching circuit conductive and to cause thegate coupling circuit to conduct the gate current to thus render thethyristor conductive at a firing time during a half cycle of the ACpower source. The control circuit renders the controllable switchingcircuit non-conductive before the end of the half-cycle, such that thegate coupling circuit is not able to conduct the gate current throughthe gate of the thyristor.

In addition, a load control circuit for controlling the amount of powerdelivered from an AC power source to an electrical load to a desiredamount of power is also described herein. The load control circuitcomprises a thyristor having a gate for conducting a gate current torender the thyristor conductive, a gate coupling circuit coupled toconduct the gate current through the gate of the thyristor, and acontrollable switching circuit coupled between the gate coupling circuitand the gate of the thyristor for conducting the gate current when thecontrollable switching circuit is conductive. The controllable switchingcircuit is rendered conductive and the gate coupling circuit is renderedconductive to conduct the gate current to thus render the thyristorconductive at a firing time during a half cycle of the AC power source.The gate coupling circuit is maintained conductive, such that the gatecoupling circuit is able to conduct the gate current again after thefiring time during the half-cycle. The controllable switching circuit isrendered non-conductive before the end of the half-cycle, such that thegate coupling circuit is not able to conduct the gate current throughthe gate of the thyristor.

According to another aspect of the present invention, a load controldevice for controlling the amount of power delivered from an AC powersource to an electrical load comprises a thyristor having a gate forconducting a gate current to render the thyristor conductive, and a gatecoupling circuit comprising two MOS-gated transistors operativelycoupled in anti-series connection between the first main load terminaland the gate of thyristor for conducting the gate current through thegate of the thyristor when the anti-series combination of the MOS-gatedtransistors is conductive. The load control device further comprises acontrol circuit operable to cause the anti-series combination of theMOS-gated transistors to conduct the gate current to thus render thethyristor conductive at a firing time during a half cycle of the ACpower source. The control circuit is operable to cause the anti-seriescombination of the MOS-gated transistors to remain conductive to conductthe gate current again after the firing time and before the end of thehalf-cycle.

According to another embodiment of the present invention, a load controldevice for controlling the amount of power delivered from an AC powersource to an electrical load comprises a thyristor having a gate forconducting a gate current to render the thyristor conductive andcharacterized by a rated holding current, and a gate coupling circuitcoupled to conduct the gate current through the gate of the thyristor,and to conduct the load current if the magnitude of the load currentfalls below the rated holding current of the thyristor. The load controldevice further comprises a control circuit operable to cause the gatecoupling circuit to conduct the gate current to thus render thethyristor conductive at a firing time during a half cycle of the ACpower source. The control circuit continues to control the gate couplingcircuit, such that the gate coupling circuit is able to conduct the gatecurrent again after the firing time.

According to another aspect of the present invention, a two-wire loadcontrol device for controlling the amount of power delivered from an ACpower source to an electrical load comprises a bidirectionalsemiconductor switch adapted to be coupled in series electricalconnection between the AC power source and the electrical load forconducting a load current from the AC power source to the electricalload, an actuator for receiving a user input, and a control circuitoperable to operate the load control device in a low-power mode inresponse to an actuation of the actuator, where the control circuitdisables one or more electrical circuits of the load control device inthe low-power mode. The bidirectional semiconductor switch is operableto be rendered conductive and to remain conductive independent of themagnitude of the load current conducted through the semiconductorswitch. The control circuit is coupled so as to conduct a controlcurrent through the electrical load and to generate a drive signal forcontrolling the bidirectional semiconductor switch to be conductive andnon-conductive in response to actuations of the actuator.

Other features and advantages of the present invention will becomeapparent from the following description of the invention that refers tothe accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will now be described in greater detail in the followingdetailed description with reference to the drawings in which:

FIG. 1 is a simplified block diagram of a lighting control systemincluding a two-wire analog dimmer switch for controlling the intensityof an LED light source according to a first embodiment of the presentinvention;

FIG. 2 is a simplified block diagram of the dimmer switch of FIG. 1according to the first embodiment of the present invention;

FIGS. 3A and 3B show example waveforms illustrating the operation of thedimmer switch of FIG. 1 according to the first embodiment of the presentinvention;

FIG. 4 is a simplified schematic diagram of the dimmer switch of FIG. 2according to the first embodiment of the present invention;

FIG. 5 is a simplified schematic diagram of a timing circuit of thedimmer switch of FIG. 2;

FIG. 6 is a simplified schematic diagram of a dimmer switch according toa second embodiment of the present invention;

FIG. 7 shows example waveforms illustrating the operation of the dimmerswitch of FIG. 6 according to the second embodiment of the presentinvention;

FIG. 8 is a simplified schematic diagram of a dimmer switch according toa third embodiment of the present invention;

FIG. 9 is a simplified block diagram of a reverse-phase control dimmerswitch according to a fourth embodiment of the present invention;

FIG. 10 is a simplified timing diagram showing examples of waveformsillustrating the operation of the dimmer switch of FIG. 9 according tothe fourth embodiment of the present invention;

FIG. 11 is a simplified schematic diagram of the dimmer switch of FIG. 9according to the fourth embodiment of the present invention;

FIG. 12 is a simplified schematic diagram of a dimmer switch accordingto an alternate embodiment of the present invention;

FIG. 13 is a simplified schematic diagram of a dimmer switch accordingto a fifth embodiment of the present invention;

FIG. 14 is a simplified timing diagram showing examples of waveformsillustrating the operation of the dimmer switch of FIG. 13 according tothe fifth embodiment of the present invention;

FIG. 15 is a simplified schematic diagram of a dimmer switch accordingto a sixth embodiment of the present invention;

FIG. 16 is a simplified schematic diagram of a dimmer switch accordingto a seventh embodiment of the present invention;

FIG. 17 is a simplified schematic diagram of a dimmer switch accordingto an eighth embodiment of the present invention;

FIG. 18 is a simplified schematic diagram of a dimmer switch having adigital control circuit according to a ninth embodiment of the presentinvention;

FIG. 19 is a simplified flowchart of a switch procedure executed by amicroprocessor of the dimmer switch of FIG. 18 according to the ninthembodiment of the present invention;

FIG. 20 is a simplified flowchart of a control procedure periodicallyexecuted by the microprocessor of the dimmer switch of FIG. 18 accordingto the ninth embodiment of the present invention;

FIG. 21 is a simplified schematic diagram of a dimmer switch accordingto a tenth embodiment of the present invention;

FIG. 22 is a simplified schematic diagram of a portion of the dimmerswitch of FIG. 21 showing first and second gate drive circuits and acontrollable switching circuit in greater detail;

FIG. 23 shows example waveforms illustrating the operation of the dimmerswitch of FIG. 21 according to the tenth embodiment of the presentinvention; and

FIG. 24 is a simplified block diagram of a dimmer switch according to aneleventh embodiment of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

The foregoing summary, as well as the following detailed description ofthe preferred embodiments, is better understood when read in conjunctionwith the appended drawings. For the purposes of illustrating theinvention, there is shown in the drawings an embodiment that ispresently preferred, in which like numerals represent similar partsthroughout the several views of the drawings, it being understood,however, that the invention is not limited to the specific methods andinstrumentalities disclosed.

FIG. 1 is a simplified block diagram of a lighting control system 10including a “two-wire” dimmer switch 100 for controlling the amount ofpower delivered to a high-efficiency lighting load 101 including a loadregulation device, e.g., a light-emitting diode (LED) driver 102, and ahigh-efficiency light source, e.g., an LED light source 104 (or “lightengine”). The dimmer switch 100 has a hot terminal H coupled to analternating-current (AC) power source 105 for receiving an AC mains linevoltage V_(AC), and a dimmed-hot terminal DH coupled to the LED driver102. The dimmer switch 100 does not require a direct connection to theneutral side N of the AC power source 105. The dimmer switch 100generates a phase-control voltage V_(PC) (e.g., a dimmed-hot voltage) atthe dimmed-hot terminal DH and conducts a load current I_(LOAD) throughthe LED driver 102. The dimmer switch 100 may either use forwardphase-control dimming or reverse phase-control dimming techniques togenerate the phase-control voltage V_(PC).

As defined herein, a “two-wire” dimmer switch or load control devicedoes not require a require a direct connection to the neutral side N ofthe AC power source 105. In other words, all currents conducted by thetwo-wire dimmer switch must also be conducted through the load. Atwo-wire dimmer switch may have only two terminals (i.e., the hotterminal H and the dimmed hot terminal DH as shown in FIG. 1).Alternatively, a two-wire dimmer switch (as defined herein) couldcomprise a three-way dimmer switch that may be used in a three-waylighting system and has at least three load terminals, but does notrequire a neutral connection. In addition, a two-wire dimmer switch maycomprise an additional connection that provides for communication with aremote control device (for remotely controlling the dimmer switch), butdoes not require the dimmer switch to be directly connected to neutral.

The LED driver 102 and the LED light source 104 may be both includedtogether in a single enclosure, for example, having a screw-in baseadapted to be coupled to a standard Edison socket. When the LED driver102 is included with the LED light source 104 in the single enclosure,the LED driver only has two electrical connections: to the dimmer switch100 for receiving the phase-control voltage V_(PC) and to the neutralside N of the AC power source 105. The LED driver 102 comprises arectifier bridge circuit 106 that receives the phase-control voltageV_(PC) and generates a bus voltage V_(BUS) across a bus capacitorC_(BUS). The LED driver 102 further comprises a load control circuit 107that receives the bus voltage V_(BUS) and controls the intensity of theLED light source 104 in response to the phase-control signal V_(PC).Specifically, the load control circuit 107 of the LED driver 102 isoperable to turn the LED light source 104 on and off and to adjust theintensity of the LED light source to a target intensity L_(TRGT) (i.e.,a desired intensity) in response to the phase-control signal V_(PC). Thetarget intensity L_(TRGT) may range between a low-end intensityL_(LE)(e.g., approximately 1%) and a high-end intensity L_(HE) (e.g.,approximately 100%). The LED driver 102 may also comprise a filternetwork 108 for preventing noise generated by the load control circuit107 from being conducted on the AC mains wiring. Since the LED driver102 comprises the bus capacitor C_(BUS) and the filter network 108, theLED driver may have a capacitive input impedance. An example of the LEDdriver 102 is described in greater detail in U.S. patent applicationSer. No. 12/813,908, filed Jun. 11, 2009, entitled LOAD CONTROL DEVICEFOR A LIGHT-EMITTING DIODE LIGHT SOURCE, the entire disclosure of whichis hereby incorporated by reference.

In addition, the LED driver 102 may comprise an artificial load circuit109 for conducting current (in addition to the load current I_(LOAD))through the dimmer switch 100. Accordingly, if the dimmer switch 100includes a triac for generating the phase-control voltage V_(PC), theartificial load circuit 109 may conduct enough current to ensure thatthe magnitude of the total current conducted through the triac of thedimmer switch 100 exceeds the rated latching and holding currents of thetriac. In addition, the artificial load circuit 109 may conduct a timingcurrent if the dimmer switch 100 comprises a timing circuit and mayconduct a charging current if the dimmer switch comprises a powersupply, such that these currents need not be conducted through the loadcontrol circuit 107 and do not affect the intensity of the LED lightsource 104.

The artificial load circuit 109 may simply comprise a constant impedancecircuit (e.g., a resistor) or may comprise a current source circuit.Alternatively, the artificial load circuit 109 may be controllable, suchthat the artificial load circuit may be enabled and disabled to thusselectively conduct current through the dimmer switch 100. In addition,the artificial load circuit 109 may be controlled to conduct differentamounts of current depending upon the magnitude of the AC mains linevoltage V_(AC), the present time during a half cycle of the AC mainsline voltage, or the present operating mode of the LED driver 102.Examples of artificial load circuits are described in greater detail incommonly-assigned U.S. patent application Ser. No. 12/438,587, filedAug. 5, 2009, entitled VARIABLE LOAD CIRCUITS FOR USE WITH LIGHTINGCONTROL DEVICES, and U.S. patent application Ser. No. 12/950,079, filedNov. 19, 2010, entitled CONTROLLABLE-LOAD CIRCUIT FOR USE WITH A LOADCONTROL DEVICE, the entire disclosures of which are hereby incorporatedby reference.

Alternatively, the high-efficiency light source could comprise a compactfluorescent lamp (CFL) and the load regulation device could comprise anelectronic dimming ballast. In addition, the dimmer switch 100 couldalternatively control the amount of power delivered to other types ofelectrical loads, for example, by directly controlling a lighting loador a motor load. An example of a screw-in light source having afluorescent lamp and an electronic dimming ballast is described ingreater detail in U.S. patent application Ser. No. 12/704,781, filedFeb. 12, 2010, entitled HYBRID LIGHT SOURCE, the entire disclosure ofwhich is hereby incorporated by reference.

The dimmer switch 100 comprises a user interface having a rocker switch116 and an intensity adjustment actuator 118 (e.g., a slider knob asshown in FIG. 1). The rocker switch 116 allows for turning on and offthe LED light source 104, while the intensity adjustment actuator 118allows for adjustment of the target intensity L_(TRGT) of the LED lightsource 104 from the low-end intensity L_(LE) to the high-end intensityL_(HE). Examples of user interfaces of dimmer switches are described ingreater detail in commonly-assigned U.S. patent application Ser. No.12/363,258, filed Jan. 30, 2009, entitled LOAD CONTROL DEVICE HAVING AVISUAL INDICATION OF ENERGY SAVINGS AND USAGE INFORMATION, the entiredisclosure of which is hereby incorporated by reference.

FIG. 2 is a simplified block diagram of the dimmer switch 100 accordingto a first embodiment of the present invention. FIGS. 3A and 3B showexample waveforms illustrating the operation of the dimmer switch 100according to the first embodiment of the present invention. The dimmerswitch 100 comprises a bidirectional semiconductor switch 110 coupledbetween the hot terminal H and the dimmed hot terminal DH for generatingthe phase-control voltage V_(PC) (as shown in FIGS. 3A and 3B) andcontrolling of the amount of power delivered to the LED driver 102. Thebidirectional semiconductor switch 110 comprises a control input (e.g.,a gate), which may receive control signals for rendering thebidirectional semiconductor switch conductive and non-conductive. Thebidirectional semiconductor switch 110 may comprise a single device,such as a triac, or a combination of devices, such as, two field-effecttransistors (FETs) coupled in anti-series connection. According to thefirst embodiment of the present invention, the phase-control voltageV_(PC) comprises a forward phase-control voltage. In other words, thephase-control voltage V_(PC) has a magnitude of approximately zero voltsat the beginning of each half cycle during a non-conduction time T_(NC),and has a magnitude equal to approximately the magnitude of the AC linevoltage V_(AC) of the AC power source 105 during the rest of the halfcycle, i.e., during a conduction time T_(CON). For example, theconduction time T_(CON) may be approximately two milliseconds when thetarget intensity L_(TRGT) of the LED light source 104 is at the low-endintensity L_(LE) and approximately seven milliseconds when the targetintensity L_(TRGT) is at the high-end intensity L_(HE).

The dimmer switch 100 comprises a mechanical air-gap switch S112electrically coupled to the hot terminal H and in series with thebidirectional semiconductor switch 110, such that the LED light source104 is turned off when the switch is open. When the air-gap switch S112is closed, the dimmer switch 100 is operable to control thebidirectional semiconductor switch 110 to control the amount of powerdelivered to the LED driver 102. The air-gap switch S112 is mechanicallycoupled to the rocker switch 116 of the user interface of the dimmerswitch 100, such that the switch may be opened and closed in response toactuations of the rocker switch. The dimmer switch 100 further comprisesa rectifier circuit 114 coupled across the bidirectional semiconductorswitch 110 and operable to generate a rectified voltage V_(RECT) (i.e.,a signal representative of the voltage developed across thebidirectional semiconductor switch).

According to the first embodiment, the dimmer switch 100 comprises ananalog control circuit 115 including a power supply 120, a constant-rateone-shot timing circuit 130, and a variable-threshold trigger circuit140 (i.e., a gate drive circuit). The control circuit 115 receives therectified voltage V_(RECT) from the rectifier circuit 114 and conducts acontrol current I_(CNTL) through the load (i.e., the LED driver 102) inorder to generate a drive voltage V_(DR) for controlling thebidirectional semiconductor switch 110 to thus adjust the intensity ofthe LED light source 104 in response to the intensity adjustmentactuator 118. The power supply 120 of the control circuit 115 conducts acharging current I_(CHRG) through the LED driver 102 in order togenerate a supply voltage V_(CC) (e.g., approximately 11.4 volts). Thecharging current I_(CHRG) of the power supply makes up a portion of thecontrol current I_(CNTL) of the control circuit 115.

The timing circuit 130 receives the supply voltage V_(CC) and generatesa timing voltage V_(TIM) (i.e., a timing signal), which comprises a rampsignal having a constant rate of increasing magnitude (i.e., a constantpositive slope) as shown in FIGS. 3A and 3B. When the bidirectionalsemiconductor switch 110 is non-conductive at the beginning of each halfcycle, the timing circuit 130 also receives the rectified voltageV_(RECT) and is able to derive zero-crossing timing information from thevoltage developed across the LED driver 102 (i.e., from the controlcurrent I_(CNTL) conducted through the LED driver 102). The timingvoltage V_(TIM) begins increasing from approximately zero volts shortlyafter the zero-crossings of the AC line voltage V_(AC) (i.e., shortlyafter the beginning of each half cycle as shown at times t₁, t₄ in FIGS.3A and 3B) and continues increasing at the constant rate. After a fixedamount of time T_(TIM) has elapsed since the timing voltage V_(TIM)started increasing from zero volts during the present half cycle, thetiming voltage V_(TIM) is driven to approximately zero volts near thenext zero-crossing (i.e., near the end of the present half cycle asshown at time t₃ in FIGS. 3A and 3B). Since the timing voltage V_(TIM)increases in magnitude at the constant rate for the fixed amount of timeT_(TIM) each half cycle, the timing voltage V_(TIM) is essentiallyidentical during each half cycle as shown in FIGS. 3A and 3B.

Referring back to FIG. 2, the variable-threshold trigger circuit 140receives the timing voltage V_(TIM) from the timing circuit 130, andgenerates a drive voltage V_(DR) (i.e., a gate drive voltage) forcontrolling the bidirectional semiconductor switch 110 to thus adjustthe intensity of the LED light source 104 in response to actuations ofthe intensity adjustment actuator 118. The trigger circuit 140 ischaracterized by a variable threshold (i.e., a variable thresholdvoltage V_(TH) shown in FIGS. 3A and 3B) that may be adjusted inresponse to the intensity adjustment actuator 118 of the user interfaceof the dimmer switch 100.

A gate coupling circuit 150 couples the drive voltage V_(DR) to the gateof the bidirectional semiconductor switch 110 for thus rendering thebidirectional semiconductor switch 110 conductive and non-conductive inresponse to the magnitude of the variable threshold voltage V_(TH). Whenthe magnitude of the timing voltage V_(TIM) exceeds the magnitude of avariable threshold voltage V_(TH) each half cycle (as shown at firingtimes t₂, t₅ in FIGS. 3A and 3B), the trigger circuit 140 is operable todrive the drive signal V_(DR) to a first magnitude (e.g., approximatelyzero volts as shown in FIGS. 3A and 3B) to thus render the bidirectionalsemiconductor switch 110 conductive each half cycle (as will bedescribed in greater detail below with reference to FIG. 4). The drivesignal V_(DR) is then driven to a second magnitude (e.g., approximatelythe supply voltage V_(CC) as shown in FIGS. 3A and 3B) to render thebidirectional semiconductor switch 110 non-conductive when the timingvoltage V_(TIM) is controlled to approximately zero volts shortly beforethe next zero-crossing. The variable threshold voltage V_(TH) is shownat two different magnitudes in FIGS. 3A and 3B, which results in thedrive signal V_(DR) being driven low to zero volts (and thus renderingthe bidirectional semiconductor switch 110 conductive) for differentamounts of time.

As shown in FIGS. 3A and 3B, the control circuit 115 of the dimmerswitch 100 is operable to provide a constant gate drive to thebidirectional semiconductor switch 110 by maintaining the drive voltageV_(DR) low for the remainder of the half cycle after the bidirectionalsemiconductor switch 110 is rendered conductive (as shown at firingtimes t₂, t₅). Accordingly, the bidirectional semiconductor switch 110will remain conductive independent of the magnitude of the load currentI_(LOAD) conducted through the bidirectional semiconductor switch andthe LED driver 102. When the bidirectional semiconductor switch 110 isconductive and the magnitude of the phase control voltage V_(PC) isgreater than approximately the magnitude of the bus voltage V_(BUS) ofthe LED driver 102, the LED driver 102 will begin to conduct the loadcurrent I_(LOAD) through the bidirectional semiconductor switch. Sincethe bus capacitor C_(BUS) of the LED driver 102 may charge quickly, themagnitude of the load current I_(LOAD) may quickly peak before subsidingdown to a substantially small magnitude (e.g., approximately zero amps).As previously mentioned, the bidirectional semiconductor switch 110 willremain conductive independent of the magnitude of the load currentI_(LOAD) because the control circuit 115 is providing constant gatedrive to the bidirectional semiconductor switch. In addition to quicklyincreasing and decreasing in magnitude, the load current I_(LOAD) mayalso change direction after the bidirectional semiconductor switch 110is rendered conductive. Therefore, the bidirectional semiconductorswitch 110 is also operable to conduct current in both directions (i.e.,to and from the LED driver 102) after the bidirectional semiconductorswitch is rendered conductive during a single half cycle, therebyallowing any capacitors in the filter network 108 of the LED driver 102to follow the magnitude of the AC line voltage V_(AC) of the AC powersource 105.

FIG. 4 is a simplified schematic diagram of the dimmer switch 100. Asshown in FIG. 4, the bidirectional semiconductor switch 110 of thedimmer switch 100 of the first embodiment is implemented as a triac110′, but may alternatively be implemented as one or moresilicon-controlled rectifiers (SCRs), or any suitable thyristor. Thetriac 110′ comprises two main terminals that are coupled in serieselectrical connection between the hot terminal H and the dimmed hotterminal DH, such that the triac is adapted to be coupled in serieselectrical connection between the AC power source 105 and the LED driver102 for conducting the load current I_(LOAD) to the LED driver. Thetriac 110′ comprises a gate (i.e., a control input) for rendering thetriac conductive each half cycle of the AC power source 105 as will bedescribed in greater detail below. While not shown in FIG. 4, a chokeinductor may be coupled in series with the triac 110′, and a filtercircuit (such as a filter capacitor) may be coupled between the hotterminal H and the dimmed hot terminal DH (i.e., in parallel with thetriac) to prevent noise generated by the switching of the triac frombeing conducted on the AC mains wiring.

The rectifier circuit 114 comprises a full-wave rectifier bridge havingfour diodes D114A, D114B, D114C, D114D. The rectifier bridge of therectifier circuit 114 has AC terminals coupled in series between the hotterminal H and the dimmed hot terminal DH, and DC terminals forproviding the rectified voltage V_(RECT) to the timing circuit 130 whenthe triac 110′ is non-conductive and a voltage is developed across thedimmer switch 100. The control circuit 115 conducts the control currentI_(CNTL) through the rectifier circuit 114 and the LED driver 102.Accordingly, the total current conducted through the LED driver 102 eachhalf cycle is the sum of the load current I_(LOAD) conducted through thebidirectional semiconductor switch 110, the control current I_(CNTL)conducted through the control circuit 115 of the dimmer switch 100, andany leakage current conducted through the filter circuit (that may becoupled between the hot terminal H and the dimmed hot terminal DH).

As shown in FIG. 4, the power supply 120 comprises, for example, apass-transistor circuit that generates the supply voltage V_(CC). Thepass-transistor circuit comprises an NPN bipolar junction transistorQ122 having a collector coupled to receive the rectifier voltageV_(RECT) through a resistor R124 (e.g., having a resistance ofapproximately 100 kΩ). The base of the transistor Q122 is coupled to therectifier voltage V_(RECT) through a resistor R125 (e.g., having aresistance of approximately 150 kΩ), and to circuit common through azener diode Z126 (e.g., having a break-over voltage of approximately 12volts). The power supply 120 further comprises a storage capacitor C128,which is able to charge through the transistor Q122 to a voltage equalto approximately the break-over voltage of the zener diode Z126 minusthe base-emitter drop of the transistor Q122. The storage capacitor C128has, for example, a capacitance of approximately 10 kF, and operates tomaintain the supply voltage V_(CC) at an appropriate magnitude (i.e.,approximately 11.4 volts) to allow the timing circuit 120 to generatethe timing voltage V_(TIM) and the gate coupling circuit 150 to continuerendering the triac 110′ conductive after the firing times each halfcycle.

The timing circuit 130 comprises a constant ramp circuit 160, a one-shotlatch circuit 170, and a reset circuit 180. The constant ramp circuit160 receives the supply voltage V_(CC) and causes the timing voltageV_(TIM) to increase in magnitude at the constant rate. The reset circuit180 receives the rectified voltage V_(RECT) and is coupled to the timingvoltage V_(TIM), such that the reset circuit is operable to start thetiming voltage V_(TIM) increasing in magnitude from approximately zerovolts shortly after the beginning of each half cycle at a half cyclestart time (e.g., times t₁, t₄ in FIGS. 3A and 3B). Specifically, thereset circuit 180 is operable to enable the timing voltage V_(TIM)(i.e., to start the increase of the magnitude of the timing voltageV_(TIM)) in response to a positive-going transition of the rectifiedvoltage V_(RECT) across a reset threshold V_(RST) that remains above thereset threshold V_(RST) for at least a predetermined amount of time. Theone-shot latch circuit 170 provides a latch voltage V_(LATCH) to thereset circuit 180 to prevent the reset circuit 180 from resetting thetiming voltage V_(TIM) until the end of the half cycle, thus ensuringthat the reset circuit only restarts the generation of the timingvoltage once each half cycle.

The one-shot latch circuit 170 stops the generation of the timingvoltage V_(TIM) by controlling the magnitude of the timing voltageV_(TIM) to approximately 0.6 volts at the end of the fixed amount oftime from when the reset circuit 180 enabled the timing voltage V_(TIM)(e.g., near the end of the half cycle at time t₃ in FIGS. 3A and 3B).After the one-shot latch circuit 170 controls the magnitude of thetiming voltage V_(TIM) to approximately 0.6 volts, the reset circuit 180is once again able to enable the generation of the timing voltageV_(TIM) after the beginning of the next half cycle (i.e., at time t₄ inFIGS. 3A and 3B). As a result, a dead time T_(DT) exists between thetime when the one-shot latch circuit 170 drives the timing voltageV_(TIM) to approximately 0.6 volts and the reset circuit 180 enables thegeneration of the timing voltage V_(TIM) by controlling the magnitude ofthe timing voltage V_(TIM) down to approximately zero volts.

The variable-threshold trigger circuit 140 comprises a comparator U142having an inverting input that receives the timing voltage V_(TIM) fromthe timing circuit 130. The variable-threshold trigger circuit 140 alsocomprises a potentiometer R144 that is mechanically coupled to theslider knob of the intensity adjustment actuator 118. The potentiometerR144 has a resistive element coupled between the supply voltage V_(CC)and circuit common and a wiper terminal that generates the variablethreshold voltage V_(TH). The variable threshold voltage V_(TH)comprises a DC voltage that varies in magnitude in response to theposition of the slider knob of the intensity adjustment actuator 118 andis provided to a non-inverting input of the comparator U142. The drivevoltage V_(DR) is generated at an output of the comparator U142 and isprovided to the gate coupling circuit 150 for rendering the triac 110′conductive and non-conductive. The gate coupling circuit 150 comprisesan opto-coupler U152 having an input photodiode, which is coupledbetween the supply voltage V_(CC) and the output of the comparator U142and in series with a resistor R154 (e.g., having a resistance ofapproximately 8.2 kΩ). The opto-coupler U152 has an output phototriacthat is coupled in series with a resistor R156 (e.g., having aresistance of approximately 100 S2). The series combination of theoutput phototriac of the opto-coupler U152 and the resistor R156 iscoupled between the gate and one of the main terminals of the triac 110′(e.g., to the hot terminal H).

As shown in FIGS. 3A and 3B, when the magnitude of the timing voltageV_(TIM) is below the magnitude of the variable threshold voltage V_(TH),the magnitude of the drive voltage V_(DR) at the output of thecomparator U142 of the variable-threshold trigger circuit 140 remainshigh at approximately the supply voltage V_(CC), such that the triac110′ remains non-conductive. When the magnitude of the timing voltageV_(TIM) increases above the variable threshold voltage V_(TH), thecomparator U142 drives the drive voltage V_(DR) low to approximatelycircuit common, such that the input photodiode of the opto-coupler U152conducts a drive current I_(DR), which may have an rated magnitudeI_(DR-RTD) of approximately 2 mA. As a result, the output phototriac ofthe opto-coupler U152 is rendered conductive and conducts a gate currentI_(G) through the gate of the triac 110′, thus rendering the triacconductive. Accordingly, the drive voltage V_(DR) is driven low torender the triac 110′ conductive after a variable amount of time haselapsed since the half cycle start time (i.e., the non-conduction timeT_(NC) as shown in FIGS. 3A and 3B), where the variable amount of timeis adjusted in response to intensity adjustment actuator 118 and thevariable threshold voltage V_(TH). Since the magnitude of the drivevoltage V_(DR) remains low after the triac 110′ is rendered conductive,the input photodiode of the opto-coupler U152 continues to conduct thedrive current I_(DR) for the remainder of the half cycle. For example,the input photodiode of the opto-coupler U152 may conduct an averagecurrent from the storage capacitor C128 of the power supply 120 wherethe average current may range from approximately 0.5 milliamps when thetarget intensity L_(TRGT) of the LED light source 104 is at the low-endintensity L_(LE) to approximately 1.7 milliamps when the targetintensity L_(TRGT) is at the high-end intensity L_(HE).

As previously mentioned, the load current I_(LOAD) may change directionafter the triac 110′ is rendered conductive (i.e., the magnitude of theload current I_(LOAD) transitions from positive to negative or viceversa). When the magnitude of the load current I_(LOAD) falls below theholding current of the triac 110′, the triac commutates off and becomesnon-conductive. In addition, the gate of the triac 110′ stops conductingthe gate current I_(G) and the output phototriac of the opto-couplerU152 becomes non-conductive. However, because the magnitude of the drivevoltage V_(DR) remains low and accordingly, the input photodiode of theopto-coupler U152 continues to conduct the drive current I_(DR) (i.e.,providing a constant gate drive) even when the triac 110′ becomesnon-conductive, the output phototriac of the opto-coupler is able toconduct the gate current I_(G) and the triac 110′ is able to be renderedconductive and conduct the load current I_(LOAD) in the oppositedirection shortly thereafter. Accordingly, the triac 110′ is able toconduct the load current I_(LOAD) in both directions in a single halfcycle.

After the triac 110′ is rendered conductive each half cycle, the timingcircuit 130 continues to generate the timing voltage V_(TIM). Thus, themagnitude of the timing voltage V_(TIM) remains above the variablethreshold voltage V_(TH) and the triac 110′ remains conductive untilapproximately the end of the half cycle when the one-shot latch circuit170 drives the timing voltage to approximately zero volts. The inputphotodiode of the opto-coupler U152 continues to conduct the drivecurrent I_(DR) and the output phototriac continues to conduct the gatecurrent I_(G) to render the triac 110′ conductive while the drivevoltage V_(DR) is driven low each half cycle (as shown in FIGS. 3A and3B).

According to the first embodiment of the present invention, the latchcircuit 170 is operable to control the timing voltage V_(TIM) toapproximately zero volts (thus controlling the magnitude of the drivevoltage V_(DR) high to approximately the supply voltage V_(CC)) shortlybefore the end of the present half cycle (as shown at time t₃ in FIGS.3A and 3B). Accordingly, the length of the timing voltage V_(TIM) (i.e.,the fixed amount of time T_(TIM)) is slightly smaller than the lengthT_(HC) of each half cycle. The dead time T_(DT) (or “blanking pulse”) inthe timing voltage V_(TIM) at the end of the half cycle allows the triac110′ to commutate off (i.e., become non-conductive) when the magnitudeof the load current I_(LOAD) through the triac reduces to approximatelyzero amps at the end of the half cycle.

Because the LED driver 102 may have a capacitive input impedance, themagnitude of the phase-control voltage V_(PC) may not quickly decreaseto zero volts near the zero-crossing of the AC mains lines voltageV_(AC) after the triac 110′ becomes non-conductive at the end of eachhalf cycle. Therefore, according to the first embodiment of the presentinvention, the reset circuit 180 only starts the timing voltage V_(TIM)after a zero-crossing of the AC mains lines voltage V_(AC), i.e., inresponse to the magnitude of the rectified voltage V_(RECT) exceedingthe reset threshold V_(RST) when the rectified voltage is increasing inmagnitude. The reset circuit 180 is prevented from resetting the timingvoltage V_(TIM) in response to the magnitude of the rectified voltageV_(RECT) dropping below the reset threshold V_(RST), which may or maynot happen each half cycle due to the capacitive input impedance of theLED driver 102.

FIG. 5 is a simplified schematic diagram of the timing circuit 130. Theconstant ramp circuit 160 receives the supply voltage V_(CC) andgenerates the timing voltage V_(TIM) across a timing capacitor C162(e.g., having a capacitance of approximately 50 nF). The constant rampcircuit 160 comprises a constant current source for conducting aconstant timing current I_(TIM) through the timing capacitor C162, suchthat the timing voltage V_(TIM) has a constant slope. The constantcurrent source circuit comprises a PNP bipolar junction transistor Q164having an emitter coupled to the supply voltage V_(CC) via a resistorR165 (e.g. having a resistance of approximately 10 kΩ). Two diodes D166,D168 are coupled in series between the supply voltage V_(CC) and thebase of the transistor Q164. A resistor R169 is coupled between the baseof the transistor Q164 and circuit common and has, for example, aresistance of approximately 51 kΩ. A voltage having a magnitude ofapproximately the forward voltage drop of the diode D166 (e.g.,approximately 0.6 V) is produced across the resistor R165, such that theresistor conducts the constant timing current I_(TIM) (e.g.,approximately 70 μA) into the capacitor C162. The rate at which themagnitude of the timing voltage V_(TIM) increases with respect to time(i.e., dV_(TIM)/dt) is a function of the magnitude of the timing currentI_(TIM) and the capacitance C_(C162) of the capacitor C162 (i.e.,dV_(TIM)/dt=I_(TIM)/C₁₆₂), and may be equal to, for example,approximately 1.4 V/msec.

The one-shot latch circuit 170 comprises a comparator U172 having aninverting input coupled to the timing voltage V_(TIM). The timingvoltage V_(TIM) is further coupled to an output of the comparator U172via a diode D174. The one-shot latch circuit 170 includes a resistivedivider, which is coupled in series electrical connection between thesupply voltage V_(CC) and circuit common, and comprises two resistorsR175, R176 having, for example, resistances of approximately 100 kΩ and1 MΩ, respectively. The junction of the two resistors R175, R176produces a latch threshold voltage V_(TH-L), which is provided to anon-inverting input of the comparator U172. The non-inverting input ofthe comparator U172 is also coupled to the output via a resistor R178(e.g., having a resistance of approximately 1 kΩ). The latch voltageV_(LATCH) is generated at the output of the comparator U172 and isprovided to the reset circuit 180 as will be described in greater detailbelow.

The reset circuit 180 comprises a first comparator U181 having anon-inverting input that receives the rectified voltage V_(RECT) via theseries combination of a zener diode Z182 and a resistor R183 (e.g.,having a resistance of approximately 100 kΩ). The parallel combinationof a capacitor C184 (e.g., having a capacitance of approximately 1000pF) and a resistor R185 (e.g., having a resistance of approximately 20kΩ) is coupled between the non-inverting input of the comparator U181and circuit common. A zener diode Z186 (e.g., having a break-overvoltage of approximately 12 volts) clamps the magnitude of the voltageproduced between the non-inverting input of the comparator U181 andcircuit common. The reset circuit 180 further comprises a resistivedivider that has two resistors R187, R188 (e.g., having resistances ofapproximately 150 kΩ and 100 kΩ, respectively), and is coupled in serieselectrical connection between the supply voltage V_(CC) and circuitcommon. The junction of the two resistors R187, R188 produces a resetthreshold voltage V_(RST) (e.g., approximately 4.8 V), which is providedto an inverting input of the comparator U181. An output of thecomparator U181 is coupled to the supply voltage V_(CC) via a resistorR189 (e.g., having a resistance of approximately 10 kΩ).

The reset circuit 180 also comprises a second comparator U191 having anon-inverting input coupled to the threshold voltage V_(RST) and anoutput coupled to the timing voltage V_(TIM). The output of thecomparator U181 is coupled to an inverting input of the secondcomparator U191 via a capacitor C190 (e.g., having a capacitance ofapproximately 1000 pF). A resistor R192 (e.g., having a resistance ofapproximately 68 kΩ) and a diode D193 are coupled between the invertinginput of the comparator U191 and circuit common. A FET Q194 is alsocoupled between the inverting input and circuit common. The gate of theFET Q194 is pulled up towards the supply voltage V_(CC) through aresistor R195 (e.g., having a resistance of approximately 100 kΩ), andis coupled to the latch voltage V_(LATCH), such that the FET may berendered conductive and non-conductive in response to the one-shot latchcircuit 170.

When the timing voltage V_(TIM) starts out at approximately zero volts,the inverting input of the comparator U172 of the latch circuit 170 isless than the latch threshold voltage V_(TH-L)(e.g., approximately 10.5V) at the non-inverting input and the output is pulled up towards thesupply voltage V_(CC) via the resistor R195 and the diode D196 of thereset circuit 180. The magnitude of the timing voltage V_(TIM) continuesto increase at the constant rate until the magnitude of timing voltageexceeds the latch threshold voltage V_(TH-L), at which time, thecomparator U172 of the latch circuit 170 drives the output low toapproximately zero volts. At this time, the magnitude of the timingvoltage V_(TIM) is reduced to approximately the forward voltage drop ofthe diode D174 (e.g., approximately 0.6 V). Accordingly, the fixedamount of time T_(TIM) that the timing voltage V_(TIM) is generated eachhalf cycle is a function of the constant rate at which the magnitude ofthe timing voltage V_(TIM) increases with respect to time dV_(TIM)/dt(i.e., approximately 1.4 V/msec) and the magnitude of the latchthreshold voltage V_(TH-L) (i.e., approximately 10.5 V), such that thefixed amount of time T_(TIM) is approximately 7.5 msec each half cycle.After the magnitude of the timing voltage V_(TIM) has exceeded the latchthreshold voltage V_(TH-L), the latch threshold voltage V_(TH-L) isreduced to approximately 0.1 V, such that the comparator U172 continuesto drive the output low and the magnitude of the timing voltage V_(TIM)is maintained at approximately 0.6 V.

At the beginning of a half cycle, the magnitude of the rectified voltageV_(RECT) is below a break-over voltage of the zener diode Z182 of thereset circuit 180 (e.g., approximately 30 V) and the voltage at thenon-inverting input of the first comparator U181 is approximately zerovolts, such that the output of the first comparator is driven lowtowards circuit common. When the magnitude of the rectified voltageV_(RECT) exceeds approximately the break-over voltage of the zener diodeZ182, the capacitor C184 begins to charge until the magnitude of thevoltage at the non-inverting input of the first comparator U181 exceedsthe reset threshold voltage V_(RST). The output of the first comparatorU181 is then driven high towards the supply voltage V_(CC) and thecapacitor C190 conducts a pulse of current into the resistor R192, suchthat the magnitude of the voltage at the inverting input of the secondcomparator U191 exceeds the reset threshold voltage V_(RST), and thesecond comparator pulls the timing voltage V_(TIM) down towards circuitcommon (i.e., the magnitude of the timing voltage is controlled fromapproximately 0.6 volts to zero volts). The magnitude of the voltage atthe inverting input of the comparator U172 of the latch circuit 170 isnow less than the latch threshold voltage V_(TH-L) (i.e., approximately0.1 V), and the comparator stops pulling the timing voltage V_(TIM) downtowards circuit common. In addition, the reset circuit 180 only drivesthe timing voltage V_(TIM) low for a brief period of time (e.g.,approximately 68 μsec) before the capacitor C190 fully charges and thenstops conducting the pulse of current into the resistor R192.Accordingly, the second comparator U191 then stops pulling the timingvoltage V_(TIM) down towards circuit common, thus allowing the timingvoltage to once again begin increasing in magnitude with respect to timeat the constant rate.

After the reset circuit 180 resets the generation of the timing voltageV_(TIM) after the beginning of each half cycle, the comparator U172 ofthe latch circuit 170 stops pulling the timing voltage V_(TIM) downtowards circuit common and the magnitude of the latch voltage V_(LATCH)is pulled high towards the supply voltage V_(CC) via the resistor R195and the diode D196. At this time, the FET Q194 is rendered conductive,thus maintaining the inverting input of the second comparator U191 lessthan the reset threshold voltage V_(RST). The FET Q194 is renderednon-conductive when the comparator U172 of the one-shot latch circuit170 pulls the timing voltage V_(TIM) low near the end of the half cycle.Thus, the FET Q194 is rendered conductive for most of each half cycleand prevents the reset circuit 180 from resetting the generation of thetiming voltage V_(TIM) until after the latch circuit 170 ceases thegeneration of the timing voltage, thereby greatly improving the noiseimmunity of the dimmer switch 100 with respect to impulse noise on theAC line voltage V_(AC).

When the magnitude of the voltage at the non-inverting input of thefirst comparator U181 of the reset circuit 170 exceeds the resetthreshold voltage V_(RST), the output is then driven high towards thesupply voltage V_(CC) and the capacitor C190 charges. The FET Q194 isthen rendered conductive, and the capacitor C190 remains charged. Whenthe magnitude of the rectified voltage V_(RECT) drops below thebreak-over voltage of the zener diode Z182 at the end of each half cycleand the magnitude of the voltage at the non-inverting input of the firstcomparator U181 drops below the reset threshold voltage V_(RST), thecapacitor C190 discharges through the diode D193 and the output of thefirst comparator U181. However, the magnitude of the voltage at theinverting input of the second comparator U191 remains less than thereset threshold voltage V_(RST), and thus the reset circuit 180 does notreset the generation of the timing voltage V_(TIM) until the magnitudeof the voltage at the non-inverting input of the first comparator U181of the reset circuit 170 rises above the reset threshold voltage V_(RST)at the beginning of the next half cycle.

Accordingly, the control circuit 115 of the dimmer switch 100 of thefirst embodiment of the present invention conducts a control currentthrough the LED driver 102 and provides constant gate drive to thebidirectional semiconductor switch 110 after the bidirectionalsemiconductor switch is rendered conductive. The control circuit 115 isoperable to derive zero-crossing timing information from the voltagedeveloped across the LED driver 102, and thus from the control currentI_(CNTL) conducted through the LED driver 102. The average magnitude ofthe control current I_(CNTL) conducted through the LED driver 102 isapproximately equal to the sum of the average magnitude of the timingcurrent I_(TIM) and the drive current I_(DR), as well as the othercurrents drawn by the timing circuit 130 and the trigger circuit 140.The control circuit 115 is operable to render the bidirectionalsemiconductor switch 110 conductive each half cycle in response to thevariable threshold that is representative of the desired intensity ofthe LED light source 104 and to maintain the bidirectional semiconductorswitch conductive until approximately the end of the present half cycle.As a result, the conduction time T_(CON) of the drive voltage V_(DR)generated by the trigger circuit 140 has a length that is not dependentupon the length of the fixed amount of time T_(TIM) that the timingcircuit 130 generates the timing signal V_(TIM).

FIG. 6 is a simplified schematic diagram of a dimmer switch 200according to a second embodiment of the present invention. FIG. 7 showsexample waveforms illustrating the operation of the dimmer switch 200according to the second embodiment of the present invention. Thebidirectional semiconductor switch of the dimmer switch 200 of thesecond embodiment is implemented as two individual MOS-gatedtransistors, e.g., FETs Q210A, Q210B, coupled in anti-series connectionbetween the hot terminal H and the dimmed hot terminal DH for control ofthe amount of power delivered to the LED driver 102. The sources of theFETs Q210A, Q210B are coupled together at circuit common. The FETsQ210A, Q210B may comprise metal-oxide semiconductor FETs (MOSFETs) ormay alternatively be replaced by any suitable voltage-controlledsemiconductor switches, such as, for example, insulated gate bipolarjunction transistors (IGBTs). The FETs Q210A, Q210B have control inputs(i.e., gates) that are coupled to a gate coupling circuit 250, whichcomprises respective gate resistors R252, R254 (e.g., each having aresistance of approximately 47Ω) for coupling to the gates of the FETs adrive voltage V_(DR-INV). The drive voltage V_(DR-INV) as shown in FIG.7 is the inverse of the drive voltage V_(DR) of the first embodiment.Each FET Q210A, Q210B is rendered conductive when the voltage at thegates of the FET is driven to a rated gate threshold voltage (e.g.,approximately 10 volts). The FETs Q210A, Q210B are simultaneouslycontrolled to be conductive and non-conductive using the forwardphase-control technique, and are operable to be rendered conductive andto remain conductive independent of the magnitude of the load currentI_(LOAD) conducted through the FETs.

The dimmer switch 200 comprises a full-wave rectifier bridge thatincludes the body diodes of the two FETs Q210A, Q210B in addition to twodiodes D214A, D214B. The timing circuit 130 of the dimmer switch 200 ofthe second embodiment operates in the same manner as in the firstembodiment. The dimmer switch 200 comprises an analog control circuit215 having a variable-threshold trigger circuit 240 that is similar tothe variable-threshold trigger circuit 140 of the first embodiment.However, the trigger circuit 240 of the second embodiment comprises acomparator U242 having a non-inverting input that receives the timingvoltage V_(TIM) and an inverting input that receives a variablethreshold voltage V_(TH) from a potentiometer R244. The trigger circuit240 operates to drive the drive voltage V_(DR-INV) high towards thesupply voltage V_(CC) to render the FETs Q210A, Q210B conductive, andlow towards circuit common to render the FETs non-conductive (as shownin FIG. 7).

As shown in FIG. 7, the gates of the FETs Q210A, Q210B only conduct asmall pulse of drive current I_(DR-INV) from the power supply 120 whenthe FETs Q210A, Q210B are rendered conductive, i.e., due to the chargingof the input capacitances of the gates of the FETs (which each may have,for example, an input capacitance of approximately 100 pF). Since thedrive current I_(DR-INV) is conducted from the storage capacitor C128 ofthe power supply 120, the average magnitude of the control currentI_(CNTL) conducted through the LED driver 102 by the analog controlcircuit 215 of the dimmer switch 200 of the second embodiment is lessthan the average magnitude of the control current I_(CNTL) conducted bythe analog control circuit 115 of the dimmer switch 100 of the firstembodiment (which conducts the drive current I_(DR) through the inputphotodiode of the opto-coupler U152 for the entire time that the triac110′ is rendered conductive).

In addition, the dimmer switch 200 of the second embodiment does notrequire the opto-coupler U152 of the first embodiment, which istypically expensive and is also characterized by a rated turn-on time(e.g., approximately 35 microseconds). In the event that the loadcurrent I_(LOAD) changes direction after the triac 110′ is renderedconductive, the rated turn-on time of the opto-coupler U152 limits howquickly the triac 110′ can be rendered conductive after becomingnon-conductive. Specifically, during the time from when the triac 110′becomes momentarily non-conductive and is once again renderedconductive, the magnitude of the phase-control voltage V_(PC) across theLED driver 102 decreases while the magnitude of the voltage across thedimmer switch 100 increases. This change in the voltage across the inputof the LED driver 102 (or electronic ballast) may result in fluctuationsin the intensity of the LED light source 104 (or fluorescent lamp) forsome high-efficiency lighting loads. Because the bidirectionalsemiconductor switch of the dimmer switch 200 is implemented as FETsQ210A, Q210B and because the FETs Q210A, Q210B are operable to remainconductive independent of the magnitude of the load current, potentialfluctuations in the intensity of some high-efficiency lighting loads areavoided.

FIG. 8 is a simplified schematic diagram of a dimmer switch 300according to a third embodiment of the present invention. The dimmerswitch 300 of the third embodiment comprises the triac 110′ (as in thefirst embodiment). However, the dimmer switch 300 includes a gatecoupling circuit 350 that comprises a voltage-controlled controllablyconductive device, such as two MOS-gated transistors (e.g., FETs Q352A,Q352B) coupled in anti-series connection between the gate and a firstone of the main load terminals of the triac 110′ (e.g., the hot terminalH of the dimmer switch). The FETs Q352A, Q352B may comprise MOSFETs ormay alternatively be replaced by any suitable voltage-controlledsemiconductor switches, such as, for example, IGBTs. The sources of theFETs Q352A, Q352B are coupled together through two source resistorsR353, R354 (e.g., each having a resistance of approximately 10Ω), wherethe junction of the two resistors R353, R354 is coupled to circuitcommon. The source resistors R353, R354 operate to limit the magnitudeof the gate current I_(G) conducted through the gate of the triac 110′to a maximum gate current (e.g., approximately 0.6 amp). The gates ofthe FETs Q352A, Q352B are coupled to respective gate resistors R355,R356 (e.g., each having a resistance of approximately 47Ω). The drivevoltage V_(DR-INV) generated by the analog control circuit 215 isreceived at a control input of the gate coupling circuit 350 (i.e., thejunction of the gate resistors R355, R356).

The dimmer switch 300 comprises a resistor R358, which has a resistanceof, for example, approximately 30.9Ω and is coupled between the gate anda second one of the main load terminals of the triac 110′ (e.g., to thedimmed hot terminal DH of the dimmer switch). The dimmer switch 300further comprises a full-wave rectifier bridge that includes the bodydiodes of the FETs Q352A, Q352B and the diodes D214A, D214B, andgenerates the rectified voltage V_(RECT) that is received by the powersupply 120 and the timing circuit 130 of the control circuit 215.Accordingly, the control circuit 215 is coupled to the first main loadterminal of the triac 110′ through the body diode of the FET Q352A andthe diode D214A, and to the second main load terminal of the triacthrough the body diode of the FET Q352B, the diode D214B, and theresistor R358. Alternatively, the control circuit 215 could be directlycoupled to at least one of the main load terminals of the triac 110′, orelectrically coupled to at least one of the main load terminals of thetriac through one or more resistors.

The timing circuit 130 of the control circuit 215 generates the timingvoltage V_(TIM) and the variable-threshold trigger circuit 240 generatesthe drive voltage V_(DR-INV) as in the second embodiment (as shown inFIG. 7). When the drive voltage V_(DR-INV) is driven low towards circuitcommon, the FETs Q352A, Q352B are non-conductive, such that the triac110′ is also non-conductive. When the trigger circuit 240 drives thedrive voltage V_(DR-INV) high towards the supply voltage V_(CC) at thefiring time each half cycle, the FETs Q352A, Q352B are able to conductthe gate current I_(G) through the gate of the triac 110′ to render thetriac conductive. The drive voltage V_(DR-INV) is driven low slightlybefore the end of the half cycle, such that the blanking pulse exists atthe end of the half cycle to allow the triac 110′ to commutate off.Since the drive voltage V_(DR-INV) remains high until approximately theend of the half-cycle, the FETs Q352A, Q352B remain conductive such thatthe FETs Q352A, Q352B are able to conduct the gate current I_(G) at anytime from the firing time through approximately the remainder of thehalf cycle. Accordingly, the triac 110′ is rendered conductive from thefiring time to approximately the end of the half cycle, thereby allowingthe load current I_(LOAD) to be either polarity (i.e., positive ornegative) in any given half cycle, which is particularly important whenthe LED driver 102 has a capacitive impedance and causes the loadcurrent to change polarity before one of the zero-crossings.

The control input of the gate coupling circuit 350 only conducts thesmall pulses of drive current I_(DR-INV) from the power supply 120 whenthe FETs Q352A, Q352B are rendered conductive due to the charging of theinput capacitances of the gates of the FETs (i.e., as shown in FIG. 7).Thus, the gate coupling circuit 350 allows the analog control circuit215 to render the triac 110′ conductive and maintain the triacconductive without the need to conduct the drive current I_(DR-INV)through the control input of the gate coupling circuit duringapproximately the remainder of the half cycle (e.g., in contrast to theinput photodiode of the optocoupler U152 of the first embodimentconducting the drive current I_(DR) as shown in FIGS. 3A and 3B).Accordingly, the average magnitude of the control current I_(CNTL)conducted by the analog control circuit 215 of the dimmer switch 300 ofthe third embodiment to render the triac 110′ conductive is less thanthe average magnitude of the control current I_(CNTL) conducted by theanalog control circuit 115 of the dimmer switch 100 of the firstembodiment to render the triac 110′ conductive. For example, if the FETsQ352A, Q352B are each characterized by a turn-on time of approximatelytwo microseconds, an input impedance of approximately 100 pF, and a gatethreshold voltage of approximately 10 volts, the gate coupling circuit350 may conduct an average current of approximately 240 nanoamps fromthe storage capacitor C128 of the power supply 120 (independent of thetarget intensity L_(TRGT) of the LED light source 104).

In addition, the dimmer switch 300 of the third embodiment does notrequire the opto-coupler U152 to render the triac 110′ conductive. Aspreviously mentioned, the opto-coupler U152 is typically expensive andis characterized by the rated turn-on time, which limits how quickly thetriac 110′ can be rendered conductive after becoming non-conductive inresponse to the load current I_(LOAD) changing directions.

Since the magnitude of the gate current I_(G) conducted by the FETsQ352A, Q352B of the gate coupling circuit 350 is much less than themagnitude of the load current I_(LOAD) conducted by the triac 110′, theFETs Q352A, Q352B of the third embodiment may be sized smaller in powerrating (and accordingly, in physical size) than the FETs Q210A, Q210B ofthe dimmer switch 200 of the second embodiment (which conduct the loadcurrent I_(LOAD)). In other words, because the FETs Q352A, Q352B of thethird embodiment do not conduct the load current I_(LOAD), the FETs neednot be power devices, but can rather be signal-level devices. Therefore,the dimmer switch 300 of the third embodiment only requires one powerdevice (i.e., the triac 110′) rather than two power devices (i.e., theFETs Q210A, Q210B), which leads to lower total cost of the dimmer switch300, as well as fewer constraints to physically fit and heat sink twopower devices in a single wall-mounted load control device. In addition,the triac 110′ typically has better peak current capabilities in asingle package as compared to the two FETs Q210A, Q210B having similarsized packages.

Accordingly, the triac 110′ and the gate coupling circuit 350 of thedimmer switch 300 of the third embodiment provide a thyristor-based loadcontrol circuit that requires substantially no net average current to beconducted through the control input after the triac is renderedconductive through the remainder of the half-cycle using a constant gatedrive signal. As used herein, “substantially no net average current” isdefined as an amount of current appropriate to charge the inputcapacitances of the gates of the FETs Q352A, Q352B (or other suitableswitching devices) of the gate coupling circuit 350, for example, lessthan approximately one microamp.

FIG. 9 is a simplified block diagram of a reverse-phase control dimmerswitch 400 according to a fourth embodiment of the present invention. Asshown in FIG. 9, the bidirectional semiconductor switch 110 isimplemented as two FETs Q210A, Q210B coupled in anti-series connection(as in the second embodiment). The dimmer switch 100 comprises an analogcontrol circuit including a voltage reference circuit 420, a timingcircuit 430, and a gate drive circuit 440. The voltage reference circuit420 includes a pass-transistor circuit 460 and a snap-on circuit 470,and operates to generate a reference voltage V_(REF) (e.g.,approximately 14.4 volts) from the rectified voltage V_(RECT). Thetiming circuit 430 receives the reference voltage V_(REF) and generatesa timing voltage V_(TIM), which is representative of the targetintensity L_(TRGT) of the LED light source 104. The gate drive circuit440 generates a gate voltage V_(G), which is coupled to the gates of theFETs Q210A, Q210B via the gate coupling circuit 250 for simultaneouslyrendering the FETs conductive and non-conductive. According to thefourth embodiment of the present invention, the phase-control voltageV_(PC) generated by the dimmer switch 400 comprises a reversephase-control voltage. Accordingly, the gate drive circuit 440 operatesto render the FETs Q210A, Q210B conductive at the beginning of each halfcycle, and non-conductive at some time during each half cycle inresponse to the timing voltage V_(TIM).

FIG. 10 is a simplified timing diagram showing examples of thephase-control voltage V_(PC) generated by the dimmer switch 400, thetiming voltage V_(TIM), and the gate voltage V_(G) for driving the FETsQ210A, Q210B according to the fourth embodiment of the presentinvention. The phase-control voltage V_(PC) has a magnitude equal toapproximately the magnitude of the AC line voltage V_(AC) of the ACpower source 105 at the beginning of each half cycle during a conductiontime T_(CON), and has a magnitude of approximately zero volts during therest of the half cycle, i.e., during a non-conduction time T_(NC). Togenerate the phase-control voltage V_(PC), the gate drive circuit 440drives the gate voltage V_(G) high towards the reference voltage V_(REF)at the beginning of each half cycle, such that the FETs Q210A, Q210B arerendered conductive (as shown at time t₁ in FIG. 10). At this time, thetiming circuit 430 begins generating the timing voltage V_(TIM), whichcomprises a ramp voltage that increases in magnitude with respect totime at a rate representative of the target intensity L_(TRGT) of theLED light source 104 (i.e., in response to the intensity adjustmentactuator 118). When the magnitude of the timing voltage V_(TIM) reachesa maximum timing voltage threshold V_(T-MAX) (e.g., approximately 7.5volts), the gate drive circuit 440 renders the FETs Q210A, Q210Bnon-conductive (as shown at time t₂ in FIG. 10). The rate of the timingvoltage V_(TIM) is inversely proportional to the target intensityL_(TRGT), i.e., the rate of the timing voltage V_(TIM) increases as thetarget intensity L_(TRGT) decreases, and decreases as the targetintensity L_(TRGT) increases. After the FETs Q210A, Q210B are renderednon-conductive, the gate drive circuit 440 will render the FETsconductive once again at the beginning of the next half cycle (as shownat time t₃ in FIG. 10).

FIG. 11 is a simplified schematic diagram of the dimmer switch 400according to the fourth embodiment of the present invention. As shown inFIG. 11, the pass-transistor circuit 460 comprises an NPN bipolarjunction transistor Q462 having a collector coupled to receive therectifier voltage V_(RECT) through a resistor R464 (e.g., having aresistance of approximately 180Ω). The base of the transistor Q462 iscoupled to the rectifier voltage V_(RE)CT through a resistor R465 (e.g.,having a resistance of approximately 470 kΩ), and to circuit commonthrough a zener diode Z466 (e.g., having a break-over voltage ofapproximately 15 volts). The pass-transistor circuit 460 furthercomprises a storage capacitor C468, which is able to charge through thetransistor Q462 and a diode D469 to a voltage equal to approximately thebreak-over voltage of the zener diode Z466 minus the base-emitter dropof the transistor Q462 and the forward drop of the diode D469. Thestorage capacitor C468 has, for example, a capacitance of approximately22 μF, and operates to maintain the reference voltage V_(REF) at anappropriate magnitude (e.g., at least approximately 12 volts) to controlthe FETs Q210A, Q210B to be conductive (i.e., when there isapproximately zero volts generated across the dimmer switch 100) as willbe described in greater detail below.

The snap-on circuit 470 is coupled to the storage capacitor Q468 andcomprises a PNP bipolar junction transistor Q472. The base of thetransistor Q472 is coupled to circuit common through the seriescombination of a resistor R474 (e.g., having a resistance ofapproximately 22 kΩ) and a zener diode Z476 (e.g., having a break-overvoltage of approximately 12 volts). The reference voltage V_(REF) isgenerated across a capacitor C478, which is coupled between thecollector of the transistor Q472 and circuit common and has, forexample, a capacitance of approximately 0.1 μF. The snap-on circuit 470operates such that the reference voltage V_(REF) is only provided acrossthe capacitor C478 when the magnitude of the voltage across the storagecapacitor C468 of the pass-transistor circuit 460 exceeds the break-overvoltage of the zener diode Z476 plus the emitter-base drop of thetransistor Q472.

The timing circuit 430 receives the reference voltage V_(REF) andgenerates the timing voltage V_(TIM) across a timing capacitor C432(e.g., having a capacitance of approximately 10 nF). The timing circuit430 includes a constant current source circuit for charging thecapacitor C432 at a constant rate to generate the timing voltageV_(TIM). The constant current source circuit comprises a PNP bipolarjunction transistor Q434 having an emitter coupled to the referencevoltage V_(REF) via a resistor R435 (e.g. having a resistance ofapproximately 180 kΩ). A voltage divider circuit comprising apotentiometer R436 and two resistors R438, R439 is coupled between thereference voltage V_(REF) and circuit common. For example, thepotentiometer R436 may have a resistance ranging from approximately 0 to500 kΩ, while the resistors R438, R439 may have resistances ofapproximately 100 kΩ and 82 kΩ, respectively. The junction of thepotentiometer R436 and the resistor R438 is coupled to the base of thetransistor Q434. The resistance of the potentiometer R436 varies inresponse to the intensity adjustment actuator 118 of the dimmer switch100, such that the magnitude of the voltage at the base of thetransistor Q434 is representative of the target intensity L_(TRGT). Whenthe potentiometer R436 is not presently being adjusted (i.e., is in asteady state condition), a constant voltage is generated across theresistor R435 and the emitter-base junction of the transistor Q434, suchthat the transistor Q434 conducts a constant current (having a magnitudedependent upon the magnitude of the voltage at the base of thetransistor Q434). Accordingly, the capacitor C432 charges at a ratedependent upon the target intensity L_(TRGT) thus generating the timingvoltage V_(TIM) (as shown in FIG. 10).

The gate drive circuit 440 renders the FETs Q210A, Q210B conductive atthe beginning of each half cycle, and non-conductive at some time duringeach half cycle in response to the timing voltage V_(TIM) from thetiming circuit 430. The gate drive circuit 440 comprises an NPN bipolarjunction transistor Q441 and a resistor R442, which is coupled betweenthe collector and base of the transistor Q441 and has a resistance of,for example, approximately 270 kΩ. A diode D443 is coupled between theemitter and the base of the transistor Q441. At the beginning of eachhalf cycle, the resistor R442 conducts current into the base of thetransistor Q441. The transistor Q441 is thus rendered conductive and thereference voltage V_(REF) is coupled to the gates of the FETs Q210A,Q210B via the respective gate resistors R252, R254 to thus render theFETs conductive. As previously mentioned, the storage capacitor C468 ofthe voltage reference circuit 420 maintains the reference voltageV_(REF) at an appropriate magnitude (i.e., at least approximately 14.4volts) to maintain the FETs Q210A, Q210B conductive and the voltagedeveloped across the dimmer switch 400 is approximately zero volts.

The timing voltage V_(TIM) is coupled to the base of an NPN bipolarjunction transistor Q444 through a zener diode Z445 (e.g., having abreak-over voltage of approximately 6.8 volts). When the magnitude ofthe timing voltage V_(TIM) exceeds approximately the break-over voltageof the zener diode Z445 plus the base-emitter drop of the transistorQ444 (i.e., the maximum timing voltage threshold V_(T-MAX)), thetransistor Q444 is rendered conductive. Accordingly, the gate voltageV_(G) is pulled down towards circuit common through the diode D443 thusrendering the FETs Q210A, Q210B non-conductive.

The gate drive circuit 440 also comprises an NPN bipolar junctiontransistor Q446 coupled across the zener diode Z445. The base of thetransistor Q446 is coupled to the junction of two series-connectedresistors R447, R448 (e.g., having resistances of approximately 200 kΩand 10 kΩ respectively). The resistors R447, R448 form a voltage dividercoupled between the rectified voltage V_(RECT) and circuit common. Thebase of the transistor Q446 is also coupled to circuit common via acapacitor C449 (e.g., having a capacitance of approximately 10 nF). Whenthe FETs Q210A, Q210B are rendered non-conductive (in response to thetiming voltage V_(TIM) exceeding the maximum timing voltage thresholdV_(T-MAX)), the voltage developed across the dimmer switch 400 increasesto approximately the magnitude of the AC line voltage V_(AC) of the ACpower source 105. As a result, the voltage at the base of the transistorQ446 increases such that the transistor is rendered conductive.Accordingly, the magnitude of the timing voltage V_(TIM) is controlledto approximately zero volts and the transistor Q444 is maintainedconductive (thus keeping the FETs Q210A, Q210B non-conductive) until theend of the present half cycle.

Near the end of the half cycle, the magnitude of the AC line voltageV_(AC) of the AC power source 105 as well as the magnitude of voltage atthe base of the transistor Q446 decrease such that the transistor Q446is rendered non-conductive. Accordingly, the transistor Q444 is renderednon-conductive and the reference voltage V_(REF) is coupled to the gatesof the FETs Q210A, Q210B through the transistor Q441 and the respectivegate resistors R252, R254, thus rendering the FETs conductive. Inaddition, when the transistor Q446 is non-conductive, the timing voltageV_(TIM) of the timing circuit 430 may once again begin increasing inmagnitude with respect to time at the rate dependent upon the targetintensity L_(TRGT) (as shown in FIG. 10).

FIG. 12 is a simplified schematic diagram of a dimmer switch 480according to an alternate embodiment of the present invention. Thedimmer switch 480 of FIG. 12 is very similar to the dimmer switch 400 ofthe fourth embodiment. However, the dimmer switch 480 of FIG. 12comprises a voltage compensation circuit 490, which receives therectified voltage V_(RECT) and adjusts the timing voltage V_(TIM) toaccount for changes and fluctuations in the AC line voltage V_(AC) ofthe AC power source 105 to avoid flickering of the intensity of the LEDlight source 104. The voltage compensation circuit 490 comprises tworesistors R492, R494, which are coupled in series between the rectifiedvoltage V_(RECT) and circuit common, and have, for example, resistancesof approximately 1 MΩ and 98 kΩ, respectively. A capacitor C496 iscoupled between the junction of the resistors R492, R494 and circuitcommon, and has, for example, a capacitance of approximately 0.22 μF.The capacitor C496 is coupled to the timing voltage V_(TIM) through aresistor R498 (e.g., having a resistance of approximately 560 kΩ).

The voltage produced across the capacitor C496 is proportional to themagnitude of the AC line voltage V_(AC) of the AC power source 105 whenthe FETs Q210A, Q210B are non-conductive and the timing voltage V_(TIM)is increasing in magnitude with respect to time. When there are nochanges or fluctuations in the magnitude of the AC line voltage V_(AC)of the AC power source 105, the capacitor C496 charges to a steady-statevoltage. However, if the magnitude of the AC line voltage V_(AC) changeswhile the FETs Q210A, Q210B are non-conductive during a half cycle(e.g., between times t₂ and t₃ in FIG. 10), the magnitude of the voltageacross the capacitor C496 will also change, thus resulting in a changein the timing voltage V_(TIM) when the FETs are conductive during thenext half cycle (e.g., between times t₃ and t₄). For example, if themagnitude of the AC line voltage V_(AC) (and thus the magnitude of thevoltage across the capacitor C496) increases while the FETs Q210A, Q210Bare non-conductive during a half cycle, the magnitude of the timingvoltage V_(TIM) will be greater while the FETs are conductive during thenext half cycle, thus causing the FETs to be rendered non-conductiveearlier in the next half cycle.

FIG. 13 is a simplified schematic diagram of a dimmer switch 500according to a fifth embodiment of the present invention. The dimmerswitch 500 comprises a mechanical air-gap switch S514 and two FETsQ510A, Q510B coupled in anti-series connection between the hot terminalH and the dimmed hot terminal DH for generating the phase-controlvoltage V_(PC). The dimmer switch 500 comprises an analog controlcircuit (e.g., a timing circuit 520) for generating a timing voltageV_(TIM) representative of the target intensity L_(TRGT) of the LED lightsource 104, and a gate drive circuit 530 for rendering the FETs Q510A,Q510B conductive and non-conductive in response to the timing voltageV_(TIM) to thus generate the phase-control voltage V_(PC). According tothe fifth embodiment of the present invention, the gate drive circuit530 is operable to generate two gate voltages V_(G1), V_(G2) forindependently controlling the respective FETs Q510A, Q510B on acomplementary basis. The FETs Q510A, Q510B are rendered conductive whenthe magnitudes of the respective gate voltages V_(G1), V_(G2) arecontrolled to a nominal gate voltage V_(N) (e.g., approximately 9 V) andare rendered non-conductive when the magnitudes of the respective gatevoltages V_(G1), V_(G2) are controlled to approximately zero volts. Thedimmer switch 500 further comprises an overcurrent protection circuit540 for rendering the FETs Q510A, Q510B non-conductive in the event ofan overcurrent condition in the FETs.

FIG. 14 is a simplified timing diagram showing examples of thephase-control voltage V_(PC) generated by the dimmer switch 500 and thegate voltages V_(G1), V_(G2) for driving the FETs Q510A, Q510B,respectively. According to the fifth embodiment of the presentinvention, the phase-control voltage V_(PC) comprises a forwardphase-control voltage. During the positive half cycles, the first FETQ510A is rendered conductive and the second FET Q510B is renderednon-conductive when the first gate voltage V_(G1) increases fromapproximately zero volts to the nominal gate voltage V_(N) (as shown attime t₁), and the second gate voltage V_(G2) decreases from the nominalgate voltage V_(N) to approximately zero volts. At this time, the dimmerswitch 500 conducts the load current I_(LOAD) to the LED driver 102through the first FET Q510A and the body diode of the second FET Q510B.At the beginning of the negative half cycles, the first FET Q510 remainsconductive. However, since the second FET Q510B is non-conductive andthe body diode of the second FET Q510B is reversed-biased, the dimmerswitch 500 does not conduct the load current I_(LOAD) at this time.

During the negative half cycles, the first FET Q510A is renderednon-conductive and the second FET Q510B is rendered conductive when thefirst gate voltage V_(G1) decreases from the nominal gate voltage V_(N)to approximately zero volts and the second gate voltage V_(G2) increasesfrom approximately zero volts to the nominal gate voltage V_(N) (asshown at time t₂). At this time, the dimmer switch 500 conducts the loadcurrent I_(LOAD) to the LED driver 102 through the second FET Q510B andthe body diode of the first FET Q510A. At the beginning of the positivehalf cycles, the second FET Q510B remains conductive, the first FETQ510A remains non-conductive, and the body diode of the first FET Q510Ais reversed-biased at this time, such that the dimmer switch 500 doesnot conduct the load current I_(LOAD) until the first FET Q510A isrendered conductive.

The timing circuit 520 is coupled in series between the hot terminal Hand the dimmed hot terminal DH and conducts a timing current I_(TIM)(i.e., a control current) through the LED driver 102 in order togenerate the timing voltage V_(TIM) across a capacitor C522 (e.g.,having a capacitance of approximately 0.1 μF). The capacitor C522 isoperable to charge from the AC power source 105 through resistors R524,R525 (e.g., having resistances of approximately 27 kΩ and 10 kΩ,respectively) and a potentiometer R526. The resistance of thepotentiometer R526 may range from, for example, approximately 0 kΩ to300 kΩ, and may be controlled by a user of the dimmer switch 500 (e.g.,by actuating the slider control) to adjust the target intensity L_(TRGT)of the LED light source 104. A calibration resistor R527 is coupled topotentiometer R526 for calibrating the range of the potentiometer, andhas a resistance of, for example, approximately 300 kΩ. Since thecapacitor C522 charges through the potentiometer R526, the rate at whichthe capacitor C522 charges and thus the magnitude of the timing voltageV_(TIM) are representative of the target intensity L_(TRGT) of the LEDlight source 104.

The drive circuit 530 comprises a diac 532 (e.g., having a break-overvoltage V_(BR) of approximately 32 volts) and two pulse transformers534A, 534B. The diac 532 is coupled in series with the primary windingsof the two pulse transformers 534A, 534B. The secondary windings of thepulse transformers 534A, 534B are coupled to respective capacitorsC535A, C535B via respective zener diodes Z536A, Z536B (which each have abreak-over voltage approximately equal to the nominal gate voltageV_(N), i.e., approximately 9 V). The capacitors C535A, C535B are coupledto the gates of the FETs Q510A, Q510B via gate resistors R538A, R538B,respectively (e.g., having resistances of approximately 47 kΩ). The gateresistors R538A, R538B may alternatively have different resistances inorder to change the duration of the switching times of the FETs Q510A,Q510B as is well known in the art.

When the magnitude of the timing voltage V_(TIM) exceeds approximatelythe break-over voltage V_(BR) of the diac 532, the diac conducts a pulseof current (i.e., a firing current I_(FIRE) as shown in FIG. 13) throughthe primary windings of the pulse transformers 534A, 534B causingsecondary voltages V_(SEC) (e.g., approximately 9V) to be generatedacross the secondary windings of the pulse transformers. During thepositive half cycles, the capacitor C535A charges from the secondarywinding of the first pulse transformer 534A through the zener diodeZ536A to approximately the nominal gate voltage V_(N) (i.e.,approximately 9 volts). Accordingly, the first gate voltage V_(G1) isdriven high from approximately zero volts to the nominal gate voltageV_(N) rendering the first FET Q510A conductive (as shown at time t₁ inFIG. 14). At the beginning of the negative half cycles, the first FETQ510A is conductive, while the second FET Q510B is non-conductive. Sincethe body diode of the second FET Q510B is reversed biased at this time,the dimmer switch 500 does not conduct the load current I_(LOAD).

During the negative half cycles, the firing current I_(FIRE) has anegative magnitude, thus causing the secondary voltages V_(SEC) acrossthe secondary windings of the pulse transformers 534A, 534B to also havenegative magnitudes. Accordingly, the zener diode Z536A isreverse-biased during the negative half cycles, causing the capacitorC535A to discharge through the zener diode Z536A, such that the voltageacross the capacitor C535A is driven to approximately zero volts. As aresult, the first gate voltage V_(G1) is driven low from the nominalgate voltage V_(N) to approximately zero volts rendering the first FETQ510A non-conductive (as shown at time t₂ in FIG. 14). In addition, thezener diode Z536B coupled to the secondary winding of the second pulsetransformer 534B is forward-biased in the negative half cycles, suchthat the capacitor C535B charges to approximately the nominal gatevoltage V_(N) and the second FET Q510B is rendered conductive during thenegative half cycles (as shown at time t₂ in FIG. 14). Accordingly, theFETs Q510A, Q510B are driven in a complementary manner, such that—at alltimes—at least one FET is conductive, while the other FET isnon-conductive. As a result, the FETs Q510A, Q510B are driven to beconductive for approximately the period T_(HC) of a half cycle andnon-conductive for the period T_(HC) of a half cycle.

The timing circuit 520 also comprises a diac 528 (e.g., having abreak-over voltage of approximately 64V) coupled to the potentiometerR526. The diac 528 provides voltage compensation by adjusting thevoltage provided to the potentiometer R526 to compensate for variationsin the AC line voltage V_(AC) provided by the AC power source 105. Thediac 528 has a negative impedance transfer function, such that thevoltage across the diac increases as the current through the diacdecreases. Thus, as the voltage across the dimmer switch 500 (i.e.,between the hot terminal H and the dimmed hot terminal DH) decreases,the current through the resistor R524 and the diac 528 decreases. As aresult, the voltage across the diac 528 increases, thus causing thecurrent flowing through the potentiometer R526 to increase and thefiring capacitor C522 to charge at a faster rate. This results in anincreased conduction time T_(CON) of the FETs Q510A, Q510B during thepresent half cycle to compensate for the decreased voltage across thedimmer switch 500, thereby maintaining the intensity of the LED lightsource 104 constant.

The drive circuit 530 is characterized as having inherent shorted-FETprotection. In the event that one of the FETs Q510A, Q510B failsshorted, the drive circuit 530 is operable to drive the other,non-shorted FET into full conduction, such that the load currentI_(LOAD) is not asymmetric. Asymmetric current can cause some types oflighting loads to overheat. For example, if the second FET Q510B failsshorted, the full AC waveform will be provided to the LED driver 102during the negative half cycles. Since there will be approximately zerovolts produced across the dimmer switch 500 during the negative halfcycles when second FET Q510B is shorted, the capacitor C522 of thetiming circuit 520 will not charge, the diac 532 of the drive circuit330 will not conduct the pulse of the firing current I_(FIRE), and thevoltage across the capacitor C535A will not be driven to zero volts torender the first FET Q510A non-conductive during the negative halfcycles. Accordingly, the first FET Q510A will remain conductive duringboth half cycles and the load current I_(LOAD) will be substantiallysymmetric. The second FET Q510B is controlled to be conductive in asimilar manner if the first FET Q510A has failed shorted.

The overcurrent protection circuit 540 comprises a sense resistor R542(e.g., having a resistance of approximately 0.015Ω). The sense resistorR542 is coupled between the sources of the FETs Q510A, Q510B, such thata voltage representative of the magnitude of the load current I_(LOAD)is generated across the sense resistor. The voltage generated across thesense resistor R542 is provided to the base of a first NPN bipolarjunction transistor (BJT) Q544. The first transistor Q544 is coupledacross the capacitor C535A and operates to protect the first FET Q510Ain the event of an overcurrent condition during the positive halfcycles. When the magnitude of the load current I_(LOAD) exceeds apredetermined current limit (e.g., approximately 46.6 amps) such thatthe voltage generated across the sense resistor R542 exceeds the ratedbase-emitter voltage (e.g., approximately 0.7 volts) of the firsttransistor Q544, the first transistor is rendered conductive.Accordingly, the first transistor Q544 pulls the first gate voltageV_(G1) at the gate of the first FET Q510A down towards zero volts, thusrendering the first FET non-conductive. The overcurrent protectioncircuit 540 further comprises a second NPN bipolar junction transistorQ546, which is coupled across the capacitor C535B and operates toprotect the second FET Q510B during the negative half cycles. When themagnitude of the load current I_(LOAD) exceeds the predetermined currentlimit, the second transistor Q546 is rendered conductive, thus pullingthe second gate voltage V_(G2) at the gate of the second FET Q510B downtowards zero volts and rendering the second FET non-conductive.

FIG. 15 is a simplified schematic diagram of a dimmer switch 600according to a sixth embodiment of the present invention. The dimmerswitch 600 comprises a drive limit circuit 650, which is coupled inseries with the diac 532 and the primary windings of the two pulsetransformers 534A, 534B of the drive circuit 530. The drive limitcircuit 650 operates to limit the number of times that the drive circuit530 attempts to render the FETs Q510A, Q510B conductive during aspecific half cycle. For example, if the overcurrent protection circuit540 renders one of the FETs Q510A, Q510B non-conductive, the drive limitcircuit 650 prevents the drive circuit 530 from attempting to render therespective FET conductive again during the present half cycle.

When the diac 532 fires each half cycle, the drive limit circuit 650conducts the firing current I_(FIRE) and generates an offset voltageV_(OFFSET) across a capacitor C652A during the positive half cycles anda capacitor C652B during the negative half cycles. The capacitor C452Acharges through a diode D654A during the positive half cycles, and thecapacitor C452B charges through a diode D654B during the negative halfcycles. For example, the capacitors C652A, C652B may have capacitancesof approximately 0.1 μF. Discharge resistors R656A, R656B are coupled inparallel with the capacitors C652A, C652B, respectively, and each have aresistance of, for example, approximately 33 kΩ. The drive limit circuit450 further comprises two zener diodes Z658A, Z658B coupled inanti-series connection and each having the same break-over voltage V_(Z)(e.g., approximately 40V). The zener diodes Z658A, Z658B are coupled tothe timing circuit 520 to limit the magnitude of the timing voltageV_(TIM) to a clamp voltage V_(CLAMP), i.e., approximately the break-overvoltage V_(Z), in both half cycles.

At the beginning of a positive half cycle, the capacitor C652A of thedrive limit circuit 540 has no charge, and thus, no voltage is developedacross the capacitor. The timing voltage signal V_(TIM) increases untilthe magnitude of the timing voltage V_(TIM) exceeds approximately thebreak-over voltage V_(BR) of the diac 532. When the diac 532 fires, thediode D654A and the capacitor C652A conduct pulse of the firing currentI_(FIRE) and the offset voltage V_(OFFSET) (e.g., approximately 12volts) is developed across the capacitor C652A. After the diac 532 hasfinished conducting the firing current I_(FIRE), the voltage across thecapacitor C522 decreases by approximately a break-back voltage (e.g.,approximately 10 volts) of the diac 532 to a predetermined voltage V_(P)(e.g., approximately 22 volts). If the overcurrent protection circuit540 renders one of the FETs Q510A, Q510B non-conductive, the timingvoltage signal V_(TIM) will begin to increase again. The magnitude ofthe timing voltage V_(TIM) must exceed approximately the break-overvoltage V_(BR) of the diac 532 plus the offset voltage V_(OFFSET) acrossthe capacitor C652A (i.e., approximately 44 volts) in order for the diac532 to conduct the pulse of the firing current I_(FIRE) once again.However, because the zener diode Z658A limits the timing voltage V_(TIM)to the break-over voltage V_(Z) (i.e., approximately 40 volts), thetiming voltage V_(TIM) is prevented from exceeding the voltage thresholdV_(TH). Accordingly, the drive circuit 530 is prevented from repeatedlyattempting to render the FETs Q510A, Q510B conductive during each halfcycle in the event of an overcurrent condition.

The timing voltage V_(TIM) is prevented from exceeding the voltagethreshold V_(TH) until the voltage ΔV across the capacitor C652A decaysto approximately the break-over voltage V_(Z) of the zener diode Z658Aminus the break-over voltage V_(BR) of the diac 532. The capacitor C652Adischarges slowly through the discharge resistor R656A, such that thetime required for the voltage ΔV across the capacitor C652A to decay toapproximately the break-over voltage V_(Z) of the zener diode Z658Aminus the break-over voltage V_(BR) of the diac 532 is long enough suchthat the drive circuit 530 only attempts to render the FETs Q510A, Q510Bconductive once during each half cycle. The voltage across the capacitorC652A decays to substantially zero volts during the negative half cyclesuch that the voltage across the capacitor C652A is substantially zerovolts at the beginning of the next positive half cycle. The capacitorC652B, the diode D654B, the discharge resistor R656B, and the zenerdiode Z658B of the drive limit circuit 650 operate in a similar fashionduring the negative half cycles. An example of the drive limit circuit650 is described in greater detail in commonly-assigned U.S. Pat. No.7,570,031, issued Aug. 4, 2009, entitled METHOD AND APPARATUS FORPREVENTING MULTIPLE ATTEMPTED FIRINGS OF A SEMICONDUCTOR SWITCH IN ALOAD CONTROL DEVICE, the entire disclosure of which is herebyincorporated by reference.

FIG. 16 is a simplified schematic diagram of a dimmer switch 700according to a seventh embodiment of the present invention. The dimmerswitch 700 comprises a drive circuit 730 that includes a single pulsetransformer 734. The pulse transformer 734 has a single primary windingand secondary winding having a tap connection 734′. The diac 532 iscoupled in series with the single primary winding of the pulsetransformer 734. The series combination of the zener diode Z536A and thecapacitor C535A is coupled between one end of the secondary winding andthe tap connection 734′ of the pulse transformer 734. The seriescombination of the diode Z536B and the capacitor C535B is coupledbetween the other end of the secondary winding and the tap connection734′ of the pulse transformer 734. The drive circuit 730 of the seventhembodiment operates to render the FETs Q510A, Q510B conductive andnon-conductive in the same manner as the drive circuit 530 of the fifthembodiment.

FIG. 17 is a simplified schematic diagram of a dimmer switch 800according to an eighth embodiment of the present invention. The dimmerswitch 800 comprises a mechanical air-gap switch S814 and two FETsQ810A, Q810B coupled in anti-series connection between the hot terminalH and the dimmed hot terminal DH for control of the amount of powerdelivered to the connected LED driver 102. As in the fifth, sixth, andseventh embodiments, the FETs Q810A, Q810B have control inputs (i.e.,gates) that receive respective gate voltages V_(G1), V_(G2) forrendering the FETs conductive and non-conductive. The LED light source104 is off when the switch S814 is open, and is on when the switch isclosed. The dimmer switch 800 comprises a control circuit that includesa timing circuit 820 and a power supply 880 and is operable to conduct acontrol current I_(CNTL) through the LED driver 102. The timing circuit820 conducts a timing current I_(TIM) in order to generate a timingvoltage V_(TIM) (as in the fifth embodiment). The dimmer switch 800further comprises a drive circuit 830 for rendering the FETs 810A, Q810Bconductive and non-conductive in response to the timing voltage V_(TIM)and an overcurrent protection circuit 860 for rendering the FETs 810A,Q810B non-conductive in response to an overcurrent condition through theFETs.

The power supply 880 generates a DC supply voltage V_(S) (e.g.,approximately 14.4 volts) for powering the drive circuit 830 and theovercurrent protection circuit 860. The power supply 880 conducts acharging current I_(CHRG) through the LED driver 102 when the dimmerswitch 800 is not conducting the load current I_(LOAD) to the LED driverand the magnitude of the voltage developed across the dimmer switch isapproximately equal to the magnitude of the AC line voltage V_(AC). Thecontrol current I_(CNTL) conducted through the LED driver 102 isapproximately equal to the sum of the timing current I_(TIM) of thetiming circuit 820 and the charging current I_(CHRG) of the power supply880.

The power supply 880 comprises a diode D881 coupled to the hot terminalH (via the switch S814), such that the power supply 880 only chargesduring the positive half cycles of the AC power source 105. The powersupply 880 includes a pass-transistor circuit that operates to generatethe supply voltage V_(S) across a capacitor C882 (e.g., having acapacitance of approximately 10 μF). The pass-transistor circuitcomprises an NPN bipolar junction transistor Q883, a resistor R884(e.g., having a resistance of approximately 220Ω), a resistor R885(e.g., having a resistance of approximately 470 kΩ), and a zener diodeZ886. The capacitor C882 is coupled to the emitter of the transistorQ883, such that the capacitor is able to charge through the transistor.The zener diode Z886 is coupled to the base of the transistor Q883 andhas a break-over voltage of, for example, approximately 15V, such thatthe capacitor C882 is able to charge to a voltage equal to approximatelythe break-over voltage minus the base-emitter drop of the transistor.

The power supply 880 further comprises snap-on circuit including a PNPbipolar junction transistor Q887, a resistor R888 (e.g., having aresistance of approximately 22 kΩ), and a zener diode Z889. The resistorR888 and the zener diode Z889 are coupled in series with the base of thetransistor Q887, and the collector of the transistor Q887 is coupled toa capacitor C890. The zener diode Z889 has a break-over voltage of, forexample, approximately 12 V, such that the voltage across the capacitorC882 is coupled across the capacitor C890 when the magnitude of thevoltage across the capacitor C882 exceeds approximately the break-overvoltage of the zener diode Z889 plus the emitter-base drop of thetransistor Q887. When the magnitude of the voltage across the capacitorC882 drops below approximately the break-over voltage of the zener diodeZ889 plus the emitter-base drop of the transistor Q887, the voltageacross the capacitor C882 is disconnected from the capacitor C890, suchthat the supply voltage V_(S) will drop to approximately circuit common(i.e., approximately zero volts).

The timing circuit 820 conducts the timing current I_(TIM) and generatesthe timing voltage V_(TIM) across a capacitor C822 (e.g., having acapacitance of approximately 0.047 μF). The capacitor C822 charges fromthe AC power source 105 through resistors R824, R825 (e.g., havingresistances of approximately 27 kΩ and 10 kΩ, respectively) and apotentiometer R826 (e.g., having a resistance ranging from approximately0 kΩ to 300 kΩ). A calibration potentiometer R827 is coupled across thepotentiometer R826 and has, for example, a resistance ranging fromapproximately 0 to 500 kΩ. The timing circuit 820 further comprises adiac 828, which has a break-over voltage of, for example, approximately64V, and operates to provide voltage compensation for the timing circuit(in a similar manner as the diac 528 of the timing circuit 520 of thefifth embodiment).

The drive circuit 830 generates the gate voltages V_(G1), V_(G2) forrendering the FETs Q810A, Q810B conductive and non-conductive on acomplementary basis in response to the timing voltage V_(TIM) of thetiming circuit 820. The drive circuit 830 comprises a diac 832 (e.g.,having a break-over voltage of approximately 32 volts), a resistor R834(e.g., having a resistance of approximately 680Ω), and two optocouplersU835A, U835B. When the magnitude of the timing voltage V_(TIM) exceedsapproximately the break-over voltage of the diac 832, the diac conductsa firing current I_(FIRE) through the input photodiode of the firstoptocoupler U835A during the positive half cycles, and through the inputphotodiode of the second optocoupler U835B during the negative halfcycles. Accordingly, the output phototransistor of the first optocouplerU835A is rendered conductive during the positive half cycles, and theoutput phototransistor of the second optocoupler U835B is renderedconductive during the negative half cycles. The output phototransistorsof the optocouplers U835A, U835B are between the supply voltage V_(S)and circuit common through respective resistors R836, R838, which eachhave resistances of, for example, approximately 4.7 kΩ.

The output phototransistors of the optocouplers U835A, U835B are alsocoupled to set-reset (SR) latches U840A, U840B, U840C, U840D, whichoperate to generate the gate voltages V_(G1), V_(G2) and to thus renderthe FETs Q810A, Q810B conductive and non-conductive on the complementarybasis. For example, the SR latches U840A, U840B, U840C, U840D may beimplemented as part of a single integrated circuit (IC), which may bepowered by the supply voltage V_(S). As shown in FIG. 17, the outputphototransistor of the first optocoupler U835A is coupled to the setinput of the first SR latch U840A and to the reset input of the secondSR latch U840B. The output phototransistor of the second optocouplerU835B is coupled to the set input of the second SR latch U840B and tothe reset input of the first SR latch U840A. The output of the first SRlatch U840A is coupled to the gate of the first FET Q810A and the outputof the second SR latch U840B is coupled to the gate of the second FETQ810B through respective resistors R842, R852, which each have aresistance of, for example, approximately 47 kΩ.

When the output phototransistor of the first optocoupler U835A isrendered conductive during the positive half cycles, the output of thefirst SR latch U840A is driven high towards the supply voltage V_(S)(thus rendering the first FET Q810A conductive), while the output of thesecond SR latch U840B is driven low towards circuit common (thusrendering the second FET Q810B non-conductive). Similarly, when theoutput phototransistor of the second optocoupler U835B is renderedconductive during the negative half cycles, the output of the second SRlatch U840B is driven high towards the supply voltage V_(S) (thusrendering the second FET Q810B conductive), while the output of thefirst SR latch U840A is driven low towards circuit common (thusrendering the first FET Q810A non-conductive). Since the set input ofthe first SR latch U840A is coupled to the reset input of the second SRlatch U840B, and the set input of the second SR latch is coupled to thereset input of the first SR latch, the FETs Q810A, Q810B are driven in acomplementary manner (as in the fifth embodiment), such that one of theFETs is conductive, while the other FET is non-conductive.

The overcurrent protection circuit 860 is coupled to the set inputs ofthe third and fourth SR latches U840C, U840D for rendering the FETsQ810A, Q810B non-conductive in the event of an overcurrent conditionthrough the FETs. The output of the third SR latch U840C is coupled tothe base of an NPN bipolar junction transistor Q844 via a resistor R846(e.g., having a resistance of approximately 18 kΩ). The collector of thetransistor Q844 is coupled to the gate of the first FET Q810A via aresistor R848 (e.g., having a resistance of approximately 330Ω). Thedrive circuit 830 comprises a similar circuit for coupling the output ofthe fourth SR latch U840D to the gate of the second FET Q810B.

The overcurrent protection circuit 860 comprises a sense resistor R870(e.g., having a resistance of approximately 0.015Ω). The sense resistorR870 is coupled in series between the FETs Q810A, Q810B, and circuitcommon is referenced to one side of the sense resistor (as shown in FIG.12), such that the magnitude of the voltage generated across the senseresistor is proportional to the magnitude of the load current I_(LOAD).The sense resistor R870 is coupled to the base of an NPN bipolarjunction transistor Q861 via a resistor R862 (e.g., having a resistanceof approximately 2.2 kΩ). A resistor R863 is coupled between the baseand the emitter of the transistor Q861 and has a resistance of, forexample, approximately 4.7 kΩ. The emitter of the transistor Q861 iscoupled to circuit common and the collector is coupled to the supplyvoltage V_(S) via two resistors R864, R865 (e.g., having resistances ofapproximately 18 kΩ and 4.7 kΩ, respectively). The junction of theresistors R864, R865 is coupled to the base of a PNP bipolar junctiontransistor Q866. The emitter of the transistor Q866 is coupled to thesupply voltage V_(S) and the collector is coupled to circuit commonthrough a resistor R867 (e.g., having a resistance of approximately510Ω). The collector of the transistor Q866 is coupled to the set inputof the third SR latch U840C for rendering the first FET Q810Anon-conductive in the event of overcurrent conditions during thepositive half cycles. The overcurrent protection circuit 860 comprises asimilar circuit (including transistors Q871, Q876, and resistors R872,R873, R874, R875, R877) for rendering the second FET Q810Bnon-conductive in the event of overcurrent conditions during thenegative half cycles.

In the event of an overcurrent condition during a positive half cycle,the overcurrent protection circuit 860 drives the set input of the thirdSR latch U840C high towards the supply voltage V_(S). Thus, thetransistor Q844 is rendered conductive pulling the gate voltage V_(G1)down towards circuit common and rendering the first FET Q810Anon-conductive. The output phototransistor of the second optocouplerU835B is coupled to the reset input of the third SR latch U840C, suchthat the overcurrent protection is reset during the next half cycle(i.e., the negative half cycle). Specifically, when the outputphototransistor of the second optocoupler U835B is rendered conductiveduring the negative half cycles, the reset input of the third SR latchU840C latch is driven high towards the supply voltage V_(S), thusrendering the transistor Q844 non-conductive and allowing the first SRlatch U840A to control the first FET Q810A. Similarly, the overcurrentprotection circuit 860 drives the set input of the fourth SR latch U840Dhigh towards the supply voltage V_(S), thus rendering the second FETQ810B non-conductive in the event of an overcurrent condition during anegative half cycle. The reset input of the fourth SR latch U840D isdriven high when the output phototransistor of the first optocouplerU835A is rendered conductive during the positive half cycles, thusallowing the second SR latch U840B to once again control the second FETQ810B.

FIG. 18 is a simplified schematic diagram of a “smart” dimmer switch 900that offers advanced features and functionality to a user according to aninth embodiment of the present invention. As shown in FIG. 18, thebidirectional semiconductor switch 110 of the dimmer switch 900 of theninth embodiment is implemented as the triac 110′ that is driven by thegate coupling circuit 350 having two anti-series-connected FETs Q352 a,Q352B (as in the dimmer switch 300 of the third embodiment). The dimmerswitch 900 comprises an air-gap switch S912 that may be actuated by anair-gap actuator (not shown) to allow for an air-gap disconnect betweenthe AC power source 105 and the high-efficiency lighting load 101 whenservicing the high-efficiency lighting load. The air-gap switch S912 isnot the primary means for toggling the LED light source 104 on and offas will be described in greater detail below. An example of a smartdimmer switch is described in greater detail in previously-referencedU.S. Pat. No. 5,248,919.

The dimmer switch 900 comprises a digital control circuit 915 having amicroprocessor 930 for generating a drive voltage V_(DR) (which is thesame as the drive voltage V_(DR-INV) of the third embodiment shown inFIG. 7). Alternatively, the microprocessor 930 may be implemented as amicrocontroller, a programmable logic device (PLD), an applicationspecific integrated circuit (ASIC), a field-programmable gate array(FPGA), or any suitable controller or processing device. In addition,the triac 110′ of the dimmer switch 900 could alternatively be driven bythe opto-coupler U152 of the dimmer switch 100 of the first embodiment.Further, the bidirectional semiconductor switch 110 of the dimmer switch900 of the ninth embodiment could alternatively be implemented as twoFETs in anti-series connection that are simultaneously controlled to beconductive and non-conductive (i.e., in a similar manner as the FETsQ210A, Q210B of the dimmer switch 200 of the second embodiment).

The digital control circuit 915 also comprises a power supply 920operable to conduct a charging current I_(CHRG) through the LED driver102 in order to generate a DC supply voltage V_(CC). For example, thepower supply 920 may comprise a pass-transistor circuit (as in thedimmer switch 100 of the first embodiment shown in FIG. 4) or anysuitable power supply that does not draw a large charging currentthrough the LED driver 102. The digital control circuit 915 comprises avoltage divider having two resistors R934, R935 for generating a scaledvoltage V_(SCALED) having a magnitude suitable to be provided to themicroprocessor 930. The scaled voltage V_(SCALED) is representative ofthe voltage developed across the bidirectional semiconductor switch 110.The microprocessor 930 may have an analog-to-digital converter (ADC) forsampling the scaled voltage V_(SCALED), such that the microprocessor 930is operable to determine the zero-crossings of the phase control voltageV_(PC) in response to the voltage developed across the bidirectionalsemiconductor switch 110.

The digital control circuit 915 further comprises a toggle tactileswitch S_(TOGGLE), a raise tactile switch S_(RAISE), and a lower tactileswitch S_(LOWER) for receiving user inputs. The toggle tactile switchS_(TOGGLE) may be mechanically coupled to a toggle actuator or pushbutton. The raise and lower switches S_(RAISE), S_(LOWER) may bemechanically coupled to, for example, separate raise and lower buttons,respectively, or to a rocker switch having an upper portion and a lowerportion. The toggle switch S_(TOGGLE) is coupled in series with aresistor R936 between the supply voltage V_(CC) and circuit common, andgenerates a toggle control signal V_(TOGGLE). The raise switch S_(RAISE)is coupled in series with a resistor R938 between the supply voltageV_(CC) and circuit common, and generates a raise control signalV_(RAISE). The lower switch S_(LOWER) is coupled in series with aresistor R938 between the supply voltage V_(CC) and circuit common, andgenerates a lower control signal V_(LOWER). The toggle control signalV_(TOGGLE), the raise control signal V_(RAISE), and the lower controlsignal V_(LOWER) are received by the microprocessor 930. Themicroprocessor 930 is operable to toggle the LED light source 104 on andoff in response to subsequent actuations of the toggle switchS_(TOGGLE). The microprocessor 930 is operable to increase the targetintensity L_(TRGT) of the LED light source 104 in response to actuationsof the raise switch S_(RAISE) and to decrease the target intensityL_(TRGT) in response to actuations of the lower switch S_(LOWER).Alternatively, the digital control circuit 915 could comprise apotentiometer for generating a DC voltage that is representative of thedesired intensity of the LED light source 104 and varies, for example,in magnitude in response to the position of an intensity adjustmentactuator of the dimmer switch 900 (i.e., similar to the potentiometerR144 and the intensity adjustment actuator 118 of the dimmer switch 100of the first embodiment).

In addition, the dimmer switch 900 may comprise a visual display (notshown), such as, a linear array of light-emitting diodes (LEDs), fordisplaying feedback to a user of the dimmer switch 900. For example, themicroprocessor 930 may illuminate one of the LEDs to display a visualrepresentation of the target intensity L_(TRGT) of the LED light source104. When the LED light source 104 is off, the microprocessor 930 mayilluminate the LEDs dimly to provide a nightlight feature. One of theLEDs may be illuminated to a greater intensity to display the targetintensity L_(TRGT) to which the microprocessor 930 will control the LEDlight source 104 when the LED light source is turned back on. Thenightlight feature is described in greater detail in commonly-assignedU.S. Pat. No. 5,399,940, issued Mar. 21, 1995, entitled LIGHTINGINDICATING DEVICE HAVING PLURAL ILLUMINATING ELEMENTS WITH ALL SUCHELEMENTS BEING ILLUMINATED WITH ONE BEING GREATER THAN THE OTHERS, theentire disclosure of which is hereby incorporated by reference.

Further, the microprocessor 930 of the dimmer switch 900 mayalternatively be operable to receive a digital message from a wired orwireless signal receiver. For example, the digital control circuit 915of the dimmer switch 900 may comprise a radio-frequency (RF)communication circuit (not shown), e.g., an RF transceiver, and anantenna (not shown), for transmitting and receiving RF signals. Themicroprocessor 930 may be operable to control the bidirectionalsemiconductor switch 110 in response to the digital messages receivedvia the RF signals. The microprocessor 930 and the RF transceiver areboth able to be put in a sleep mode (i.e., low-power mode) to conservebattery power. During the sleep mode, the RF transceiver is operable towake up periodically to sample (e.g., listen) for RF energy at asampling period T_(SAMPLE). In the event that the RF transceiver doesdetect the presence of any RF signals 106, the RF transceiver isoperable to wake up the microprocessor 930, such that the microprocessorcan begin processing the received RF signal. Each time that themicroprocessor 930 wakes up, additional power is consumed by themicroprocessor (since the microprocessor is fully powered when awake).Alternatively, the RF communication circuit of the dimmer switch 900 maysimply comprise an RF receiver or an RF transmitter for only receivingor transmitting RF signals, respectively. Examples of RF load controldevices and antennas for wall-mounted load control devices are describedin greater detail in commonly-assigned U.S. Pat. No. 5,982,103, issuedNov. 9, 1999, and U.S. Pat. No. 7,362,285, issued Apr. 22, 2008, bothentitled COMPACT RADIO FREQUENCY TRANSMITTING AND RECEIVING ANTENNA ANDCONTROL DEVICE EMPLOYING SAME, and U.S. patent application Ser. No.13/415,537, filed Mar. 8, 2012, entitled LOW-POWER RADIO-FREQUENCYRECEIVER, the entire disclosures of which are hereby incorporated byreference.

FIG. 19 is a simplified flowchart of a switch procedure 1000 executed bythe microprocessor 930 in response to an actuation of one of the raiseswitch S_(RAISE) or the lower switch S_(LOWER) at step 1010 (i.e., ifeither of the raise control signal V_(RAISE) and the lower controlsignal V_(LOWER) are pulled down to circuit common). If the raise switchS_(RAISE) is actuated at step 1012, the microprocessor 930 increases thetarget intensity L_(TRGT) of the LED light source 104 at step 1014 bydecreasing a firing time T_(FIRE) (which is approximately equal to thenon-conduction time T_(NC) shown in FIGS. 3A and 3B). If the lowerswitch S_(LOWER) is actuated at step 1016, the microprocessor 930decreases the target intensity L_(TRGT) of the LED light source 104 byincreasing the firing time T_(FIRE) at step 1018, before the buttonprocedure 1000 exits.

FIG. 20 is a simplified flowchart of a control procedure 1100periodically executed by the microprocessor 930 (e.g., every 100 μsec)to sample the scaled voltage V_(SCALED) and generate the drive voltageV_(DR). First, the microprocessor 930 samples the scaled voltageV_(SCALED) using the ADC at step 1110. At step 1112, the microprocessor930 determines if the scaled voltage V_(SCALED) is increasing inmagnitude and if the present sample is greater than the previous samplein order to detect a positive-going transition of the scaled voltageV_(SCALED) across a zero-crossing threshold. If the microprocessor 930detects a positive-going transition across the zero-crossing thresholdat step 1112 and a RESET flag is set at step 1114, the microprocessor930 clears the RESET flag at step 1116. The microprocessor 930 theninitializes a timer to zero and starts the timer increasing in valuewith respect to time at step 1118, before the control procedure 1100exits. If the RESET flag is not set at step 1114, the microprocessor 930does not restart the timer at step 1118.

If the timer is equal to the firing time T_(FIRE) at step 1120, themicroprocessor 930 drives the drive voltage V_(DR) low to approximatelycircuit common to render the bidirectional semiconductor switch 110conductive at step 1122, and the control procedure 1100 exits. If thetime is equal to a total time T_(TOTAL) at step 1124, the microprocessor930 drives the drive voltage V_(DR) high to approximately the supplyvoltage V_(CC) to render the bidirectional semiconductor switch 110non-conductive at step 1126. The total time T_(TOTAL) may be equal tothe fixed amount of time T_(TIM) that the timing circuit 130 generatesthe timing voltage V_(TIM) in the dimmer switch 100 of the firstembodiment (i.e., approximately 7.5 msec). At step 1128, themicroprocessor 930 sets the RESET flag at step 1128, and the controlprocedure 1100 exits. The RESET flag allows the microprocessor 930 toensure that the timer is not restarted until after the total timeT_(TOTAL).

FIG. 21 is a simplified schematic diagram of a dimmer switch 1200according to a tenth embodiment of the present invention. The dimmerswitch 1200 includes a gate coupling circuit 1250 having two FETsQ1252A, Q1252B coupled in anti-series connection between the gate and afirst one of the main load terminals of the triac 110′ (e.g., the hotterminal H of the dimmer switch). The sources of the FETs Q1252A, Q1252Bare coupled together through two resistors R1253, R1254 (e.g., eachhaving a resistance of approximately 5.6Ω), where the junction of thetwo resistors R1253, R1254 is coupled to circuit common.

The dimmer switch 1200 comprises a digital control circuit 1215 having amicroprocessor 1230 that is responsive to actuators 1236 (e.g., such asthe toggle tactile switch S_(TOGGLE), the raise tactile switchS_(RAISE), and the lower tactile switch S_(LOWER) of the ninthembodiment). The digital control circuit 1215 comprises a zero-crossdetect circuit 1234 that generates a zero-cross voltage V_(ZC) that isrepresentative of the zero-crossings of the AC line voltage. The digitalcontrol circuit 1215 also comprises a power supply 1220 operable toconduct a charging current I_(CHRG) through the LED driver 102 forgenerating a first DC supply voltage V_(CC1) (e.g., approximately 8volts) for driving the FETs Q1252A, Q1252B and a second DC supplyvoltage V_(CC2) (e.g., approximately 4 volts) for powering themicroprocessor 1230. Both of the first and second DC supply voltagesV_(CC1), V_(CC2) are referenced to circuit common and the power supply1220 conducts the charging current I_(CHRG) through circuit common. Forexample, the power supply 1220 may comprise a resistor-zener powersupply for generating the first DC supply voltage V_(CC1) and ahigh-efficiency switching power supply for generating the second DCsupply voltage V_(CC2).

The gate coupling circuit 1250 of the tenth embodiment is very similarto the gate coupling circuit 350 of the third embodiment. However, thegate coupling circuit 1250 of the tenth embodiment comprises first andsecond gate drive circuits 1260, 1270 that allow for independent controlthe FETs Q1252A, Q1252B. The microprocessor 1230 generates two drivevoltages V_(DR1), V_(DR2) that are received by the respective gate drivecircuits 1260, 1270 for rendering the respective FETs Q1252A, Q1252Bconductive and non-conductive, such that the triac 110′ may be renderedconductive to conduct the load current I_(LOAD) to the LED driver 102.The dimmer switch 1200 comprises a resistor R1258, which has aresistance of, for example, approximately 90.9Ω and is coupled betweenthe gate and one of the main load terminals of the triac 110′ (e.g., tothe dimmed hot terminal DH of the dimmer switch).

In addition, the dimmer switch 1200 comprises a controllable switchingcircuit 1280 coupled in series with the anti-series-connected FETsQ1252A, Q1252B and the gate of the triac 110′. The microprocessor 1230generates a switch control voltage V_(SW) for rendering the controllableswitching circuit 1280 conductive and non-conductive. When thecontrollable switching circuit 1280 is conductive, the FETs Q1252A,Q1252B are able to conduct a gate current I_(G) through the gate of thetriac 110′ to render the triac conductive. The microprocessor 1230 isoperable to disconnect the gate of the triac 110′ from the FETs Q1252A,Q1252B before the end of each half-cycle of the AC line voltage, suchthat the triac is able to commutate off before the end of thehalf-cycle. However, the FETs Q1252A, Q1252B may conduct the loadcurrent I_(LOAD) to the LED driver 102 after the triac 110′ and beforethe end of the present half-cycle, as will be described in greaterdetail below.

FIG. 22 is a simplified schematic diagram of a portion of the dimmerswitch 1200 showing the first and second gate drive circuits 1260, 1270and the controllable switching circuit 1280 in greater detail. The firstgate drive circuit 1260 comprises an NPN bipolar junction transistorQ1261 having a base that receives the first drive voltage V_(DR1) via aresistor R1262 (e.g., having a resistance of approximately 200 kΩ). Thecollector of the transistor Q1261 is coupled to the first DC supplyvoltage V_(CC1) through a resistor R1263 (e.g., having a resistance ofapproximately 200 kΩ), and to the base of another NPN bipolar junctiontransistor Q1264. The collector-emitter junction of the transistor Q1264is coupled in series with a diode D1265 and the collector-emitterjunction of another NPN bipolar junction transistor Q1266. The base ofthe transistor Q1266 is coupled to the first DC supply voltage V_(CC1)through a resistor R1267 (e.g., having a resistance of approximately 200kΩ) and to the collector of the transistor Q1266. The junction of thetransistor Q1266 and the diode D1265 is coupled to the gate of the firstFET Q1252A through a gate resistor R1268 (e.g., having a resistance ofapproximately 47Ω).

When the first drive voltage V_(DR1) is low (i.e., at approximatelycircuit common), the transistor Q1261 is non-conductive, such that thebase of the transistor Q1265 is pulled up towards the first DC supplyvoltage V_(CC1). Accordingly, the transistor Q1265 is renderedconductive, pulling the base of the transistor Q1266 and the gate of thefirst FET Q1252A down towards circuit common, such that the FET isnon-conductive. However, when the first drive voltage V_(DR1) is high(i.e., at approximately the first DC supply voltage V_(CC1)), thetransistor Q1261 becomes conductive, such that the transistor Q1265 isrendered non-conductive. Thus, the transistor Q1266 becomes conductiveand the gate of the first FET Q1252A is driven up towards the first DCsupply voltage V_(CC1), such that the FET is rendered conductive. Thesecond gate drive circuit 1270 has an identical structure and operationfor rendering the second FET Q1252B conductive and non-conductive inresponse to the second drive voltage V_(DR2).

The controllable switching circuit 1280 is coupled between theanti-series-connected FETs Q1252A, Q1252B and the gate of the triac 110′and is responsive to the switch control voltage V_(SW) from themicroprocessor 1230. The gate of the triac 110′ is coupled to one of themain terminals through the parallel combination of a capacitor C1290(e.g., having a capacitance of approximately 0.1 μF) and a resistorR1292 (e.g., having a resistance of approximately 47Ω). The controllableswitching circuit 1280 includes a full-wave rectifier bridge comprisesfour diodes D1281-D1284. The AC terminals of the rectifier bridge arecoupled in series with the gate of the triac 110′, while an NPN bipolarjunction transistor Q1285 is coupled across the DC terminals of therectifier bridge. The controllable switching circuit 1280 also comprisesan optocoupler U1286 having an output phototransistor that is coupled inseries with a resistor R1287 across the DC terminals of the bridge. Forexample, the resistor R1287 may have a resistance of approximately 150kΩ. The switch control voltage V_(SW) is coupled to the input photodiodeof the optocoupler U1286 via a resistor R1288 (e.g., having a resistanceof approximately 10 kΩ). When the switch control voltage V_(SW) is low,the output phototransistor of the optocoupler U1286 is non-conductive,such that the transistor Q1285 is non-conductive (i.e., the controllableswitching circuit 1280 is non-conductive). However, when the switchcontrol-voltage V_(SW) is high, the output phototransistor of theoptocoupler U1286 is rendered conductive, such that the transistor Q1285is conductive (i.e., the controllable switching circuit 1280 isconductive and the gate of the triac 110′ is coupled to theanti-series-connected FETs Q1252A, Q1252B).

FIG. 23 shows example waveforms illustrating the operation of the dimmerswitch 1200 according to the tenth embodiment of the present invention.The microprocessor 1230 is operable to determine the zero-crossing ofthe AC line voltage at time t₁ in response to the zero-cross voltageV_(ZC) generated by the zero-cross detect circuit 1234. At the beginningof each half-cycle, the FETs Q1252A, Q1252B are rendered non-conductive,such that the first FET Q1252A blocks current during the positivehalf-cycles and the second FET Q1252B blocks current during the negativehalf-cycles. The microprocessor 1230 drives both of the drive voltagesV_(DR1), V_(DR2) high at the same time, such that the FETs Q1252A,Q1252B are operable to conduct the gate current I_(G) through the gateof the triac 110′ to render the triac conductive at the desired phaseangle (e.g., at time t₃ as shown in FIG. 23).

During the positive half-cycles, the microprocessor 1230 drives thesecond drive voltage V_(DR2) low at time t₅ before the end of thehalf-cycle (i.e., at time t₆ in FIG. 23), such that the second FETQ1252B will be non-conductive and ready to block current at thebeginning of the negative half-cycle. After the second drive voltageV_(DR2) is driven low at time t₅, the second FET Q1252B is operable toconduct current through its body diode until the end of the half-cycle.The microprocessor 1230 drives the first drive voltage V_(DR1) low afterthe end of the half-cycle at time t₇, such that the first FET Q1252Aremains conductive until the end of the present positive half-cycle.Similarly, during the negatives half-cycles, the microprocessor 1230drives the first drive voltage V_(DR1) low before the end of thehalf-cycle and drives the second drive voltage V_(DR2) low after the endof the half-cycle.

The microprocessor 1230 drives the switch control voltage V_(SW) high(e.g., at time t₂ as shown in FIG. 23) to cause the controllableswitching circuit 1280 to become conductive) prior to rendering the FETsQ1252A, Q1252B conductive, for example, approximately 40 μsec before thetime t₃ when the FETs are rendered conductive. If the FETs Q1252A,Q1252B allow the gate of the triac 110′ to conduct the gate currentI_(G) too close to the end of the half-cycle, the triac 110′ maymistakenly be rendered conductive at the beginning of the nexthalf-cycle, which could cause the triac to be conductive for the entirenext half-cycle and thus cause flicker in the LED light source 104.Therefore, the microprocessor 1230 drives the switch control voltageV_(SW) low (e.g., at time t₄ in FIG. 23) to cause the controllableswitching circuit 1280 to become non-conductive before the end of thepresent half-cycle, e.g., approximately 600-1000 μsec before the end ofthe half-cycle (i.e., at time t₆ in FIG. 23). Since the controllableswitching circuit 1280 becomes non-conductive before the end of thehalf-cycle, the triac 110′ is able to commutate off when the magnitudeof the load current I_(LOAD) drops below the rated holding current ofthe triac. The triac 110′ cannot be rendered conductive again during thepresent half-cycle and will remain non-conductive at the beginning ofthe next half-cycle. If the LED driver 102 needs to conduct currentafter the triac 110′ commutates off, the FETs Q1252A, Q1252B are able toconduct the load current I_(LOAD). Accordingly, the dimmer switch 1200is able to conduct current through the LED driver 102 independent of therated holding current of the triac 110′ and without driving the triac110′ too close to the next half-cycle.

Since the dimmer switch 1200 comprises the microprocessor 1230, thedimmer switch may offers advanced features and functionality to a userof the dimmer switch. The user may be able adjust the features andfunctionality of the dimmer switch 1200 using, for example, an advancedprogramming mode. The microprocessor 1230 may be operable to enter theadvanced programming mode in response to one or more actuations of theactuators 1236. For example, the user may adjust the low-end intensityL_(LE) and the high-end intensity L_(HE) between which themicroprocessor 1230 may control the target intensity L_(TRGT) of the LEDlight source 104. A dimmer switch having an advanced programming mode isdescribed in greater detail in commonly-assigned U.S. Pat. No.7,190,125, issued Mar. 13, 2007, entitled PROGRAMMABLE WALLBOX DIMMER,the entire disclosure of which is hereby incorporated by reference.

In addition, the user may cause the dimmer switch 1200 to enter alow-power mode using the advanced programming mode, i.e., in response toone or more actuations of the actuators 1236. In the low-power mode, themicroprocessor 1230 may disable one or more of the electrical circuitsof the dimmer switch 1200 (i.e., loads of the power supply 1220) todecrease the amount of current conducted through the LED driver 102 whenthe LED light source 104 is off. For example, the microprocessor 1230may be operable to turn off the LEDs, such that the dimmer switch 1200does not provide the nightlight feature when the LED light source 104 isoff. Further, the microprocessor 1230 may be operable to disable the RFcommunication circuit when the LED light source 104 is off.Alternatively, the microprocessor 1230 could increase the samplingperiod T_(SAMPLE), such that the RF communication circuit wakes up lessoften to sample for RF energy and thus consumes less power.

FIG. 24 is a simplified block diagram of a smart dimmer switch 1300according to an eleventh embodiment of the present invention. The dimmerswitch 1300 is very similar to the dimmer switch 1200 of the tenthembodiment. However, the dimmer switch 1300 has an earth ground terminalGND that is adapted to be coupled to earth ground. The zero-crossingdetector 1234 and the power supply 1220 of the dimmer switch 1300 arecoupled between the hot terminal H and the earth ground terminal GND(rather than the dimmed hot terminal DH). Accordingly, the power supply1220 conducts the charging current I_(CHRG) through the earth groundterminal GND (rather than the LED driver 102).

The smart dimmer switches 900, 1200, 1300 of the ninth, tenth, andeleventh embodiments could alternatively comprise analog dimmerswitches, e.g., dimmer switches having the mechanical air-gap switchS112 coupled to the rocker switch 116 for turning the LED light source104 on and off and an intensity adjustment actuator 118 for adjustingthe intensity of the LED light source 104 as in the first through eighthembodiments. The microprocessors 930, 1230 of the dimmer switches 900,1200, 1300 of the ninth, tenth, and eleventh embodiments would simply beunpowered when the mechanical air-gap switch S112 is open.

While the present invention has been described with reference to thehigh-efficiency lighting load 101 having the LED driver 102 forcontrolling the intensity of the LED light source 104, the dimmerswitches 100, 200, 300, 400, 500, 600, 700, 800, 900, 1200, 1300 couldbe used to control the amount of power delivered to other types oflighting loads (such as incandescent lamps, halogen lamps, magneticlow-voltage lamps, electronic low-voltage lamps), other types ofelectrical loads (such as motor and fan loads), and other types of loadregulation devices (such as electronic dimming ballasts for fluorescentlamps).

This application is related to commonly-assigned U.S. patent applicationSer. No. 12/953,057, filed Nov. 23, 2010, entitled TWO-WIRE ANALOGFET-BASED DIMMER SWITCH, the entire disclosure of which is herebyincorporated by reference.

Although the present invention has been described in relation toparticular embodiments thereof, many other variations and modificationsand other uses will become apparent to those skilled in the art. It ispreferred, therefore, that the present invention be limited not by thespecific disclosure herein, but only by the appended claims.

What is claimed is:
 1. A load control device for controlling powerdelivered from an AC power source to an electrical load, the loadcontrol device comprising: a bidirectional semiconductor switch adaptedto be electrically coupled between the AC power source and theelectrical load, and configured to conduct a load current to energizethe electrical load, the bidirectional semiconductor switch comprisesfirst and second switching transistors coupled in anti-seriesconnection, the bidirectional semiconductor switch having a controlinput configured to receive at least one drive signal to render thebidirectional semiconductor switch conductive and non-conductive; acontrol circuit configured to control the bidirectional semiconductorswitch to render the bidirectional semiconductor switch conductive andnon-conductive, the control circuit configured to determine azero-crossing time of a present half-cycle of the AC power source inresponse to a voltage developed across the bidirectional semiconductorswitch; and a power supply electrically coupled to conduct a chargingcurrent through the electrical load to generate a supply voltage;wherein the control circuit is configured to control the magnitude ofthe drive signal to a first magnitude to render the bidirectionalsemiconductor switch conductive after a first, variable amount of timehas elapsed since the zero-crossing time of the present half-cycle, thecontrol circuit configured to maintain the bidirectional semiconductorswitch conductive independent of the magnitude of the load currentconducted through the bidirectional semiconductor switch during thepresent half-cycle, the control circuit further configured to controlthe magnitude of the drive signal to a second magnitude to render thebidirectional semiconductor switch non-conductive after a second, fixedamount of time has elapsed since the zero-crossing time of the presenthalf-cycle, the second, fixed amount of time being less than a length ofthe present half-cycle, the control circuit configured to vary thefirst, variable amount of time in accordance with a determinate amountof power to be delivered to the electrical load in order to cause theamount of power delivered to the electrical load to be the determinateamount.
 2. The load control device of claim 1, wherein the controlcircuit comprises: a timing circuit configured to generate a timingsignal from the supply voltage, the timing circuit being configured tostart to generate the timing signal near the zero-crossing time of thepresent half-cycle, and to cause the timing signal to increase inmagnitude with respect to time; and a drive circuit arranged to receivethe timing signal, the drive circuit configured to control the magnitudeof the drive signal to the first magnitude to render the bidirectionalsemiconductor switch conductive in each half-cycle in response to thetiming signal reaching a selected threshold; wherein the timing circuitis further configured to continue generating the timing signal after thebidirectional semiconductor switch is rendered conductive in eachhalf-cycle, resulting in the control circuit maintaining thebidirectional semiconductor switch conductive independent of themagnitude of the load current conducted through the bidirectionalsemiconductor switch.
 3. The load control device of claim 2, wherein thetiming circuit is configured to stop generating the timing signal afterthe second, fixed amount of time has elapsed since the zero-crossingtime of the present half-cycle.
 4. The load control device of claim 3,wherein the drive circuit is configured to control the magnitude of thedrive signal to the second magnitude when the timing circuit stopsgenerating the timing signal to render the bidirectional semiconductorswitch non-conductive.
 5. The load control device of claim 3, whereinthe timing circuit is configured to increase the magnitude of the timingsignal at a constant rate and the length of the first, variable amountof time is not dependent upon the length of the second, fixed amount oftime.
 6. The load control device of claim 1, wherein the first andsecond switching transistors comprise first and second FETs.
 7. The loadcontrol device of claim 6, wherein the control circuit renders both thefirst and second FETs conductive and non-conductive at the same timeseach half-cycle to control the amount of power delivered to the load. 8.The load control device of claim 1, wherein a dead time exists betweenthe time at which the control circuit adjusts the drive signal to thesecond magnitude during a first half-cycle and the time at which thecontrol circuit determines a subsequent zero-crossing time in a secondhalf-cycle immediately following the first half-cycle.
 9. The loadcontrol device of claim 1, wherein the control circuit comprises amicroprocessor powered by the supply voltage.
 10. A load control devicefor controlling power delivered from an AC power source to an electricalload, the load control device comprising: a bidirectional semiconductorswitch adapted to be electrically coupled between the AC power sourceand the electrical load and configured to conduct a load current toenergize the electrical load, the bidirectional semiconductor switchcomprises first and second switching transistors coupled in anti-seriesconnection, the bidirectional semiconductor switch having a controlinput configured to receive at least one drive signal for rendering thebidirectional semiconductor switch conductive and non-conductive; acontrol circuit configured to control the bidirectional semiconductorswitch to render the bidirectional semiconductor switch conductive andnon-conductive, the control circuit configured to control the magnitudeof the drive signal to a first magnitude to render the bidirectionalsemiconductor switch conductive at a firing time during each half-cycleof the AC power source; and a power supply electrically coupled toconduct a charging current through the electrical load to generate asupply voltage; wherein the control circuit is configured to maintainthe drive signal at the first magnitude after the firing time, therebyenabling the bidirectional semiconductor switch to conduct the loadcurrent to the electrical load and to conduct the load current from theelectrical load during a single half-cycle of the AC power source. 11.The load control device of claim 10, wherein the control circuit isconfigured to determine a zero-crossing time of a present half-cycle ofthe AC power source in response to a voltage developed across thebidirectional semiconductor switch.
 12. The load control device of claim11, wherein the control circuit is configured to control the magnitudeof the drive signal to the first magnitude to render the bidirectionalsemiconductor switch conductive at the firing time when a variableamount of time has elapsed since the zero-crossing time of the presenthalf-cycle of the AC power source, and to maintain the bidirectionalsemiconductor switch conductive independent of the magnitude of the loadcurrent conducted through the bidirectional semiconductor switch betweenthe firing time and a second time during the present half-cycle, thecontrol circuit configured to control the magnitude of the drive signalto a second magnitude to render the bidirectional semiconductor switchnon-conductive after the second time which occurs when a fixed amount oftime has elapsed since the zero-crossing time of the present half-cycle,the fixed amount of time being less than a length of the presenthalf-cycle, the control circuit configured to vary the firing time inaccordance with a determinate amount of power to be delivered to theelectrical load in order to cause the amount of power delivered to theelectrical load to be the determinate amount.
 13. The load controldevice of claim 12, wherein the control circuit comprises: a timingcircuit configured to generate a timing signal from the supply voltage,the timing circuit being configured to start to generate the timingsignal near the zero-crossing time of the present half-cycle, and tocause the timing signal to increase in magnitude with respect to time;and a drive circuit arranged to receive the timing signal, the drivecircuit configured control the magnitude of the drive signal to thefirst magnitude to render the bidirectional semiconductor switchconductive in each half-cycle in response to the timing signal reachinga selected threshold; wherein the timing circuit is further configuredto continue generating the timing signal after the bidirectionalsemiconductor switch is rendered conductive in each half-cycle, thetiming circuit configured to stop generating the timing signal after thesecond, fixed amount of time has elapsed since the zero-crossing time ofthe present half-cycle, the drive circuit configured to control themagnitude of the drive signal to the second magnitude when the timingcircuit stops generating the timing signal to render the bidirectionalsemiconductor switch non-conductive.
 14. The load control device ofclaim 10, wherein the first and second switching transistors comprisefirst and second FETs.
 15. The load control device of claim 14, whereinthe control circuit renders both the first and second FETs conductiveand non-conductive at the same times each half-cycle to control theamount of power delivered to the load.
 16. The load control device ofclaim 10, wherein the control circuit is configured to maintain thebidirectional semiconductor switch conductive between the firing timeand a second time before the end of the half-cycle, and to control themagnitude of the drive signal to a second magnitude to render thebidirectional semiconductor switch non-conductive after the second time.17. The load control device of claim 10, wherein the control circuit isconfigured to vary the firing time in accordance with a determinateamount of power to be delivered to the electrical load in order to causethe amount of power delivered to the electrical load to be thedeterminate amount.
 18. The load control device of claim 10, wherein thecontrol circuit is configured to generate the drive signal forcontrolling the gate coupling circuit to maintain the bidirectionalsemiconductor switch conductive independent of the magnitude of the loadcurrent conducted through the bidirectional semiconductor switch. 19.The load control device of claim 10, wherein the power supply isconfigured to conduct the charging current through the electrical loadwhen the bidirectional semiconductor switch is non-conductive togenerate the supply voltage for powering the control circuit.
 20. Theload control device of claim 10, wherein the control circuit comprises amicroprocessor powered by the supply voltage.